High fidelity audio frequency power amplifier. UMZCH VV with microcontroller control system

Audio frequency power amplifier (UMZCH) high fidelity(BB), developed in 1989 by Nikolai Sukhov, can rightly be called legendary. In its development, a professional approach was applied, based on knowledge and experience in the field of analog circuitry. As a result, the parameters of this amplifier turned out to be so high that even today this design has not lost its relevance. This article describes a slightly improved version of the amplifier. Improvements come down to the use of a new element base and the use of a microcontroller control system.

The power amplifier (PA) is an integral part of any sound reproducing complex. Many descriptions of the design of such amplifiers are available. But in the vast majority of cases, even with very good performance, there is a complete lack of service facilities. But at the present time, when microcontrollers have become widespread, it is not difficult to create a sufficiently perfect control system. At the same time, a home-made device in terms of functional saturation may not be inferior to the best branded samples. A variant of the UMZCH VV with a microcontroller control system is shown in fig. one:

Rice. 1. The appearance of the amplifier.

The original UMZCH VV circuit has sufficient parameters so that the amplifier is not the dominant source of nonlinearity in the sound reproducing path over the entire output power range. Therefore, further improvement of the characteristics of noticeable advantages no longer gives.

At least the sound quality of different phonograms differs much more than the sound quality of amplifiers. On this topic, you can quote from the magazine "Audio": " There are audible differences in categories such as loudspeakers, microphones, LP pickups, listening rooms, studio spaces, concert halls, and especially the studio and recording equipment configurations used by different recording companies. If you want to hear subtle differences in the soundstage, compare John Eargle's Delos records to Jack Renner's Telarc records, not the preamps. Or if you want to hear subtle differences in transitions, compare dmp's jazz recordings to Chesky's jazz recordings, not two interconnects.»

Despite this fact, Hi-End lovers do not stop searching for the “right” sound, which affects, among other things, the UM. In fact, the PA is an example of a very simple linear path. The current level of development of circuitry makes it possible to provide such a device with sufficiently high parameters so that the introduced distortions become invisible. Therefore, in practice, any two modern, non-eccentrically designed PAs sound the same. On the contrary, if the UM has some special, specific sound, this only says one thing: the distortions introduced by such UM are large and clearly noticeable by ear.

This does not mean that it is very easy to design a high-quality PA. There are many subtleties, both circuitry and design plan. But all these subtleties have long been known to serious PA manufacturers, and there are usually no gross errors in the designs of modern PAs. The exception is expensive Hi-End class amplifiers, which are often designed very illiterately. Even if the distortion introduced by the PA is pleasing to the ear (as lovers of tube amplifiers claim), this has nothing to do with high fidelity of sound reproduction.

To a high-quality PA, in addition to the traditional requirements of broadband and good linearity, a number of additional additional requirements. Sometimes you can hear that home use sufficient power amplifier 20-35 watts. If we are talking about average power, then this statement is true. But a real music signal can have a peak power level that is 10 to 20 times the average. Therefore, in order to obtain an undistorted reproduction of such a signal at an average power of 20 W, it is necessary to have a PA power of the order of 200 W. Here, for example, is the output of the peer review for the amplifier described in: " The only remark was the insufficient volume of the sound of large percussion instruments, which is explained by the insufficient output power of the amplifier (120 watts peak into a 4 ohm load).»

Acoustic systems (AS) are a complex load and have a very complex pattern of impedance vs. frequency. At some frequencies, it can be 3-4 times less than the nominal value. The PA must be able to operate without distortion on such a low-resistance load. For example, if the nominal impedance of the speaker system is 4 ohms, then the PA should normally work on a load with a resistance of 1 ohm. This requires very high output currents, which must be taken into account when designing the amplifier. The described amplifier satisfies these requirements.

Recently, the topic of the optimal output impedance of an amplifier has been discussed quite often in terms of minimizing speaker distortion. However, this topic is relevant only when designing active speakers. Crossover filters for passive speakers are designed based on the fact that the signal source will have a negligible output impedance. If the PA has a high output impedance, then the frequency response of such speakers will be greatly distorted. Therefore, there is nothing else left but to provide a small output impedance for the PA.

It can be seen that the new developments of the UM are mainly on the path of cost reduction, improvement of the manufacturability of the design, increase in output power, increase in efficiency, improvement of consumer qualities. This article focuses on the service functions that are implemented thanks to the microcontroller control system.

The amplifier is made in a MIDI case, its overall dimensions are 348x180x270 mm, weight is about 20 kg. The built-in microcontroller allows you to control the amplifier using an IR remote control (shared with the preamplifier). In addition, the microcontroller measures and indicates the average and quasi-peak output power, the temperature of the radiators, implements a timer shutdown and handles emergency situations. The amplifier protection system, as well as power on and off control, are implemented with the participation of a microcontroller. The amplifier has a separate standby power supply, which allows it to be in "STANDBY" mode when the main power supplies are turned off.

The described amplifier is called NSM (National Sound Machines), model PA-9000, since the name of the device is part of its design and must be present. The implemented set of service functions in some cases may turn out to be redundant, for such situations a “minimalist” version of the amplifier (model PA-2020) has been developed, which has only a power switch and a two-color LED on the front panel, and the built-in microcontroller only controls the process of turning the power on and off, complements the protection system and provides remote control of the "STANDBY" mode.

All controls and indicators of the amplifier are located on the front panel. Its appearance and purpose of the controls are shown in Fig. 2:

Rice. 2. Front panel of the amplifier.

POWER button provides complete shutdown mains amplifier. Physically, this button disconnects only the standby power source from the network; accordingly, it can be designed for a small current. The main power sources are switched on by means of relays, the windings of which are powered by a standby source. Therefore, when the “POWER” button is turned off, all amplifier circuits are guaranteed to be de-energized.

When you turn on the "POWER" button, the amplifier is fully turned on. The power-on process is as follows: the standby source immediately turns on, as evidenced by the “DUTY” standby power on LED. After some time required to reset the microcontroller, power is turned on to external sockets and the “EXT” LED lights up. Then the “MAIN” LED lights up, and the first stage of turning on the main sources takes place. Initially, the main transformers are switched on through limiting resistors, which prevent the initial inrush current due to the discharged filter capacitors. The capacitors are gradually charged, and when the measured supply voltage reaches the set threshold, the limiting resistors are removed from the circuit. At the same time, the "OPERATION" LED lights up. If during the allotted time the supply voltage has not reached the set threshold, then the process of turning on the amplifier is interrupted and the alarm indication is turned on. If the inclusion of the main sources was successful, then the microcontroller checks the status of the protection system. In the absence of emergency situations, the microcontroller allows the load relay to be turned on and the “LOAD” LED lights up.

STANDBY button manages standby mode. A short press of the button puts the amplifier into standby mode or, conversely, turns on the amplifier. In practice, it may be necessary to turn on external sockets, leaving the PA in standby mode. This is required, for example, when listening to phonograms on stereo phones or when dubbing without sound control. External sockets can be switched on/off independently for a long time (up to sound signal) by pressing the STANDBY button. The option when the PA is turned on and the sockets are turned off does not make sense, therefore it is not implemented.

The front panel has a 4-digit digital display and 5 display control buttons. The display can operate in the following modes (Fig. 3a):

• disabled
• indication of average output power [W]
• indication of quasi-peak output power
• timer status indication [M]
• temperature display of radiators [°C]

Rice. 3. Display indication options.

PEAK button turns on the display of output power and switches between average / quasi-peak power. In the output power indication mode, “W” lights up on the display, and “PEAK” also lights up for quasi-peak power. The output power is displayed in watts with a resolution of 0.1 watts. The measurement is made by multiplying the current and voltage at the load, so the readings are valid for any allowable value of the load resistance. Holding down the PEAK button until a beep turns off the display. Turning off the display, as well as switching between different display modes, occurs smoothly (one image “flows” into another). This effect is implemented in software.

TIMER button displays Current state timer, the letter "M" lights up. The timer allows you to set the time interval after which the amplifier goes into standby mode and the external sockets are turned off. It should be noted that when using this function, other components of the complex must allow power off "on the go." For a tuner and CD player, this is usually acceptable, but for some cassette decks, when the power is turned off, the LPM may not go into the “STOP” mode. For these decks, turning off the power during playback or recording is unacceptable. However, such decks are extremely rare among branded devices. On the contrary, most decks have a "Timer" switch that has 3 positions: "Off", "Record" and "Play", which allows you to immediately turn on the playback or recording mode with a simple power supply. You can also turn off these modes by simply removing the power. The amplifier timer can be programmed for the following intervals (Fig. 3b): 5, 15, 30, 45, 60, 90 and 120 minutes. If the timer is not used, it must be set to the "OFF" state. It is in this state immediately after the power is turned on.

The timer interval is set buttons "+" and "-" in timer display mode. If the timer is enabled, the TIMER LED is always lit on the display, and turning on the timer indication shows the actual current state, i.e. how many minutes are left before shutdown. In such a situation, the interval can be extended by pressing the "+" button.

"°C" button turns on the display of the temperature of the radiators, while the symbol "°C" lights up. A separate thermometer is installed on each radiator, but the maximum temperature value is displayed on the display. The same thermometers are used for fan control and for thermal protection of the output transistors of the amplifier.

For fault indication There are two LEDs on the front panel: "FAIL LEFT" and "FAIL RIGHT". When the protection is triggered in one of the PA channels, the corresponding LED lights up, and the display shows the letter name of the cause of the accident (Fig. 3c). In this case, the amplifier goes into standby mode. The amplifier has the following types of protection:

• output stage overcurrent protection
• output DC protection
• power supply failure protection
• loss protection mains voltage
• protection against overheating of the output transistors

It reacts to the DC component of the PA output voltage, which is greater than 2 V. It protects the speakers, it is also implemented in hardware. Indicated on the display as "dcF".

Responds to a drop in the supply voltage of any arm below a specified level. A significant violation of the symmetry of the supply voltages can cause a constant component to appear at the output of the PA, which is dangerous for the AU. The display shows as "UF".

Responds to the loss of several periods of mains voltage in a row. The purpose of this protection is to disconnect the load before the supply voltage drops and the transient begins. Implemented in hardware, the microcontroller only reads its state. Displayed as "prF".

overheat protection output transistors is implemented in software, it uses information from thermometers that are installed on radiators. Indicated on the display as "tF".

UM has the ability remote control . Since it is not required a large number control buttons, the same remote control is used as for controlling the preamplifier. This remote control works in the RC-5 standard and has three buttons specifically designed to control the PA. The "STANDBY" button completely duplicates the similar button on the front panel. The "DISPLAY" button allows you to switch the display mode around the ring (Fig. 3a). Holding down the DISPLAY button until a beep turns off the display. The "MODE" button allows you to change the time interval of the timer (Fig. 3b), i.e. it replaces the "+" and "-" buttons.

On the back panel amplifier (Fig. 4) installed sockets designed to power other components of the complex. These sockets have an independent shutdown, which allows you to de-energize the entire complex from the remote control.

Rice. 4. Back panel of the amplifier.

As noted earlier, the UMZCH VV circuit of Nikolai Sukhov, which is described in. The basic principles for building a high-fidelity PA are set out in. circuit diagram amplifier main board shown in fig. 5.

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Rice. 5. Schematic diagram of the main board of the amplifier.

There have been minor changes to the amplifier compared to the original design. These changes are not fundamental and are basically a transition to a newer element base.

Changed quiescent current temperature stabilization circuit. In the original design, along with the output transistors, a transistor was installed on the radiators - a temperature sensor that set the bias voltage of the output stage. In this case, only the temperature of the output transistors was taken into account. But the temperature of the terminal transistors, due to the rather large power dissipated on them, also increased significantly during operation. Due to the fact that these transistors were mounted on small individual heatsinks, their temperature could fluctuate quite dramatically, for example, as a result of changes in power dissipation or even due to external air currents. This led to the same sharp fluctuations in the quiescent current. Yes, and any other element of the PA can get quite hot during operation, since there are heat sources in one case (radiators of output transistors, transformers, etc.). This also applies to the very first composite emitter follower transistors, which did not have heatsinks at all. As a result, the quiescent current could increase several times when the PA was heated. A solution to this problem was proposed by Alexei Belov.

Usually, for temperature stabilization of the quiescent current of the PA output stages, the following scheme is used (Fig. 6a):

Rice. 6. Scheme of temperature stabilization of the quiescent current.

The bias voltage is applied to points A and B. It is allocated on a two-terminal network, which consists of a transistor VT1 and resistors R1, R2. The initial bias voltage is set by resistor R2. Transistor VT1 is usually mounted on a common radiator with VT6, VT7. Stabilization is carried out as follows: when the transistors VT6, VT7 are heated, the base-emitter drop decreases, which, at a fixed bias voltage, leads to an increase in the quiescent current. But along with these transistors, VT1 also heats up, which causes a decrease in the voltage drop across the two-terminal network, i.e. reduction of quiescent current. The disadvantage of this scheme is that the junction temperature of the remaining transistors included in the composite emitter follower is not taken into account. To take it into account, the junction temperature of all transistors must be known. The easiest way is to make it the same. To do this, it is enough to install all the transistors included in the composite emitter follower on a common radiator. At the same time, to obtain a quiescent current that does not depend on temperature, the bias voltage of the composite emitter follower must have the same temperature coefficient as that of six p-n junctions connected in series. Approximately, we can assume that the forward voltage drop across the p-n junction linearly decreases with a coefficient K, approximately equal to 2.3 mV/°C. For a composite emitter follower, this coefficient is 6 * K. To ensure such a temperature coefficient of the bias voltage is the task of a two-terminal network, which is connected between points A and B. The two-terminal network shown in fig. 6a has a temperature coefficient of (1+R2/R1)*K. When the resistor R2 adjusts the quiescent current, the temperature coefficient also changes, which is not entirely correct. The simplest practical solution is the circuit shown in Fig. 6b. In this circuit, the temperature coefficient is (1 + R3 / R1) * K, and the initial quiescent current is set by the position of the slider of the resistor R2. The voltage drop across the resistor R2, which is shunted by a diode, can be considered almost constant. Therefore, adjusting the initial quiescent current does not affect the temperature coefficient. With such a circuit, when the PA is heated, the quiescent current changes by no more than 10-20%. In order for all transistors of a composite emitter follower to be placed on a common heatsink, they must have packages suitable for mounting on a heatsink (transistors in TO-92 packages are not suitable). Therefore, other types of transistors are used in the PA, at the same time more modern ones.

In the amplifier circuit (Fig. 5), the two-terminal temperature stabilization of the quiescent current is shunted by capacitor C12. This capacitor is optional, although it does no harm either. The fact is that between the bases of the transistors of the composite emitter follower, it is necessary to provide a bias voltage, which must be constant for the selected quiescent current and not depend on the signal being amplified. In short, the variable component of the voltage on the two-terminal network, as well as on the resistors R26 and R29 (Fig. 5), must be equal to zero. Therefore, all these elements can be shunted with capacitors. But due to the low dynamic resistance of the two-terminal network, as well as the low resistance values ​​\u200b\u200bof these resistors, the presence of shunt capacitances has a very weak effect. Therefore, these capacitances are not necessary, especially since for shunting R26 and R29 their values ​​\u200b\u200bmust be quite large (about 1 μF and 10 μF, respectively).

Output transistors The PA was replaced by transistors KT8101A, KT8102A, which have a higher cutoff frequency of the current transfer coefficient. In powerful transistors, the effect of a drop in the current transfer coefficient with an increase in the collector current is quite pronounced. This effect is extremely undesirable for the PA, since here the transistors have to work at high output currents. Modulation of the current transfer coefficient leads to a significant deterioration in the linearity of the output stage of the amplifier. To reduce the influence of this effect in the output stage, two transistors are connected in parallel (and this is the minimum that you can afford).

When transistors are connected in parallel, separate emitter resistors are used to reduce the influence of their parameter spread and equalize operating currents. For the normal operation of the overcurrent protection system, a circuit for highlighting the maximum voltage value on the diodes VD9 - VD12 (Fig. 5) has been added, since now it is necessary to remove the drop not from two, but from four emitter resistors.

Other transistors a composite emitter follower is KT850A, KT851A (TO-220 package) and KT940A, KT9115A (TO-126 package). In the circuit for stabilizing the quiescent current, a composite transistor KT973A (TO-126 package) is used.

Made and replaced OU to more modern ones. The main op amp U1 has been replaced by the AD744, which has improved performance and good linearity. Op-amp U2, which operates in the zero potential maintenance circuit at the UMZCH output, has been replaced by OP177, which has a low zero offset (no more than 15 microvolts). This made it possible to abandon the bias adjustment trimmer. It should be noted that due to the AD744 circuitry, op-amp U2 must provide an output voltage close to the supply voltage (pin 8 of the AD744 op-amp in terms of constant voltage is only two p-n junction). Therefore, not all types of precision op amps will fit. As a last resort, a pull-up resistor can be applied from the op-amp output to -15 V. The op-amp U3, which works in the AC lead impedance compensation circuit, has been replaced by the AD711. The parameters of this op-amp are not so critical, so a cheap op-amp with sufficient speed and a fairly low zero offset was chosen.

Resistor dividers R49 - R51, R52 - R54 and R47, R48 are added to the circuit, which serve to remove current and voltage signals for the power measurement circuit.

Changed implementation earth circuits. Since each channel of the amplifier is now completely assembled on a single board, there is no need for multiple ground wires that must be connected at one point on the chassis. The special PCB topology provides star-shaped ground wiring. The earth star is connected by one conductor to the common terminal of the power source. It should be noted that such a topology is only suitable for completely separate power supplies for the left and right channels.

In the original amplifier circuit, the AC feedback loop spans and relay contacts that connect the load. This measure was taken to reduce the influence of the non-linearity of the contacts. However, in this case, problems with the operation of the protection for the constant component are possible. The fact is that when the amplifier is turned on, power is supplied before the load relay is turned on. At this time, a signal may be present at the PA input, and the gain of the amplifier due to a broken feedback loop is very large. In this mode, the PA limits the signal, and the offset voltage compensation circuit is generally unable to maintain a zero DC value at the PA output. Therefore, even before the load is connected, it may be found that there is a constant component at the output of the PA, and then the protection system will work. It is very easy to eliminate this effect if you use relays with changeover contacts.

Normally-closed contacts should close the feedback loop in the same way as normally open ones. In this case, when the relay is activated, the feedback is interrupted only for a very short time, during which all relay contacts are open. During this time, the relatively inertial protection for the constant component does not have time to work. On fig. Figure 7 shows the relay switching process taken with a digital oscilloscope. As can be seen, 4 ms after the voltage is applied to the relay coil, the normally closed contacts open. Approximately 3 ms later, normally-open contacts are closed (with a noticeable bounce, which lasts about 0.7 ms). Thus, the contacts are in the "flight" for about 3 ms, it is for this time that the feedback will be broken.

Rice. 7. AJS13113 relay switching process.

Protection scheme completely redesigned (Fig. 8). Now it is placed on the main board. Thus, each channel has its own independent circuitry. This is somewhat redundant, but each main board is completely autonomous and is a complete mono amplifier. Part of the protective functions is carried out by the microcontroller, but to increase the reliability, a sufficient set of them is implemented in hardware. In principle, the amplifier board can work without a microcontroller at all. Since the PA has a separate standby power supply, the protection circuit is powered by it (+12V level). This makes the behavior of the protection circuit more predictable in the event of a failure of one of the main power sources.

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Rice. 8. Amplifier protection circuit.

Over current protection includes a trigger assembled on transistors VT3, VT4 (Fig. 5), which turns on when the transistor VT13 is opened. VT13 receives a signal from the current sensor and opens when the current reaches the value set using the tuning resistor R30. The trigger turns off the current generators VT5, VT6, which leads to the locking of all transistors of the composite emitter follower. Zero voltage at the output is maintained in this mode using resistor R27 (Fig. 5). In addition, the state of the trigger is read through the chain VD13, R63 (Fig. 8), and when it turns on, at the inputs logic element U4D is set logic low. The VT24 transistor provides an open collector output for the IOF (I Out Fail) signal, which is interrogated by the microcontroller.

DC protection implemented on transistors VT19 - VT22 and logic elements U4B, U4A. The signal from the output of the amplifier through the divider R57, R59 is fed to the low-pass filter R58C23 with a cutoff frequency of about 0.1 Hz, which selects the constant component of the signal. If a constant component of positive polarity appears, then the VT19 transistor opens, connected according to the OE circuit. He, in turn, opens the transistor VT22, and a high logic level appears at the inputs of the logic element U4B. If a constant component of negative polarity appears, then the transistor VT21, connected with ABOUT, opens. Such asymmetry is a forced measure associated with the unipolar power supply of the protection circuit. In order to increase the current transfer coefficient, the cascode switching of transistors VT21, VT20 (ON - OK) is used. Further, as in the first case, the VT22 transistor opens, etc. A transistor VT23 is connected to the output of the logic element U4A, which provides an open collector output for the DCF (DC Fail) signal.

Power failure protection contains an auxiliary rectifier (Fig. 13) VD1, VD2 (VD3, VD4), which has a smoothing filter with a very small time constant. If several periods of mains voltage drop out in a row, the output voltage of the rectifier drops, and the inputs of the logic element U4C (Fig. 8) are set to a logic low level.

Logic signals from the three protection circuits described above are fed to the "OR" element U5C, at the output of which a low logic level is formed in the event of any of the circuits being triggered. In this case, the capacitor C24 is discharged through the diode VD17, and a low logic level appears at the inputs of the logic element U5B (also at the output of U5A). This leads to the closing of the transistor VT27 and turning off the relay K1. The R69C24 chain provides some minimum power-on delay in case the microcontroller for some reason does not generate an initial delay. Transistor VT25 provides an open collector output for the OKL (OK Left) or OKR (OK Right) signal. The microcontroller can disable the relay. For this, a VT26 transistor is installed. This feature is necessary to implement software protection against overheating, software relay turn-on delay and to synchronize the operation of protection systems for the left and right channels.

Interaction of the microcontroller with the hardware protection circuit the following: when the amplifier is turned on, after the supply voltage has reached the nominal value, the microcontroller polls the readiness signals of the hardware protection OKL and OKR. All this time, turning on the relay is prohibited by the microcontroller by maintaining the ENB (Enable) signal at a high logic level. As soon as the microcontroller receives ready signals, it generates a time delay and allows the relay to turn on. During the operation of the amplifier, the microcontroller constantly monitors the ready signal. In case of loss of such a signal for one of the channels, the microcontroller removes the ENB signal, thus turning off the relay in both channels. It then polls the security status signals to identify the channel and the type of security.

overheat protection implemented entirely in software. In case of overheating of the radiators, the microcontroller removes the ENB signal, which causes the load relay to turn off. To measure the temperature, a Dallas DS1820 thermometer is attached to each of the radiators. The protection is triggered when the temperature of the radiators reaches 59.8 °C. Somewhat earlier, at a temperature of 55.0 °C, a preliminary overheating message appears on the display - the temperature of the radiators is automatically displayed. The amplifier restarts automatically when the radiators cool down to 35.0 °C. Switching on at a higher temperature of the radiators is possible only manually.

To improve the cooling conditions of the elements inside the amplifier case, a small-sized fan which is located on the back panel. A fan with a brushless DC motor with a nominal supply voltage of 12 V is used, designed to cool the computer processor. Since some noise is created during the operation of the fan, which can be noticeable during pauses, a rather complex control algorithm is used. When the temperature of the radiators reaches 45.0 °C, the fan starts to work, and when the radiators cool down to 35.0 °C, the fan turns off. When the output power is less than 2 W, the fan operation is prohibited so that its noise is not noticeable. To prevent the fan from turning on and off periodically when output power fluctuates around the threshold value, the minimum fan off time is software-limited to 10 seconds. When the temperature of the radiators is 55.0 °C and above, the fan runs without switching off, since this temperature is close to the emergency temperature. If the fan turns on while the amplifier is running, then when entering the “STANDBY” mode, if the temperature of the radiators is above 35.0 °C, the fan continues to work even at zero output power. This allows the amplifier to cool down quickly.

Power supply failure protection also implemented entirely in software. The microcontroller, using the ADC, monitors the supply voltages of both channels of the amplifier. This voltage is supplied to the processor from the main boards through resistors R55, R56 (Fig. 8).

The inclusion of the main power sources is carried out stepwise. This is necessary because the load of the rectifiers is completely discharged filter capacitors, and with a sharp turn-on, a strong current surge will occur. This surge is dangerous for the rectifier diodes and can blow fuses. Therefore, when the amplifier is turned on, the relay K2 is first closed (Fig. 12), and the transformers are connected to the network through the limiting resistors R1 and R2. At this time, the threshold for the measured supply voltages is set to ±38 V by software. If this voltage threshold is not reached within set time, the switching process is interrupted. This can occur if the current drawn by the amplifier circuit is significantly increased (the amplifier is damaged). In this case, the indication of power supply failure "UF" is switched on.

If the threshold of ±38 V is reached, then relay K3 (Fig. 12) is activated, which excludes resistors from the primary circuits of the main transformers. Then the threshold is reduced to ±20 V, and the microcontroller continues to monitor the supply voltages. If during the operation of the amplifier the supply voltage drops below ± 20 V, the protection is activated and the amplifier turns off. Lowering the threshold in normal operation is necessary so that during the "drawdown" of the supply voltage under load, false protection operation does not occur.

circuit diagram processor boards shown in fig. 9. The basis of the processor is the U1 microcontroller type AT89C51 from Atmel, which operates on clock frequency 12 MHz. To increase the reliability of the system, the U2 supervisor was used, which has a built-in watchdog timer and a power monitor. To reset the watchdog timer, a separate WD line is used, on which a periodic signal is generated by software. The program is designed in such a way that this signal will be present only if the timer interrupt handler and the main program loop are executing. V otherwise the watchdog timer will restart the microcontroller.

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Rice. 9. Schematic diagram of the processor board.

The display is connected to the processor via an 8-bit bus (XP4 - XP6 connectors). For gating the registers of the display board, the signals C0..C4 are used, which are generated by the address decoder U4. Register U3 is a latch for the low byte of the address, only bits A0, A1, A2 are used. The high byte of the address is not used at all, which allowed the P2 port to be freed up for other purposes.

When you press the control buttons, sound signals are programmatically generated. For this, the BPR line is used, to which it is connected transistor key VT1, loaded on a dynamic radiator HA1.

The main boards of the left and right channels are connected to the processor board using the XP1 and XP2 connectors, respectively. These connectors supply the processor with the IOF over-current protection and DCF output protection status signals for the DCF amplifier. These signals are common to the left and right channels, and their combination is possible thanks to the outputs of the open collector protection circuit. The OKL and OKR protection readiness signals are channel-separated so that the processor can identify the channel on which the protection circuit has been triggered. The ENB signal, which comes from the processor to the protection system, allows the load relay to turn on. This signal is common to the two channels, which automatically synchronizes the operation of the two relays.

The TRR and TRL lines are used to read the thermometers mounted on the right and left channel radiators, respectively. The temperature measured by thermometers can be displayed on the display if the corresponding display mode is enabled. The maximum temperature value of the two for the left and right channels is displayed. The measured value is also used for software implementation of overheating protection.

Additionally, the XP1 and XP2 connectors have the WUR, WIR, WUL, and WIL signals that are used by the output power measurement circuit.

The processor board is powered from the standby source via the XP3 connector. 4 levels are used for power supply: ±15 V, +12 V and +5 V. The ±15 V levels are turned off when switching to standby mode, and the remaining levels are always present. Consumption from levels +5 V and +12 V in standby mode is minimized due to software shutdown of the main consumers. In addition, several control logic signals are sent to the standby power source through this connector: PEN - controls the standby power source, REX - turns on the relay of external sockets, RP1 and RP2 - turns on the relay of the main power source, FAN - turns on the fan. The protection circuits located on the main boards are powered by the processor board at +12 V, and the display board is powered by +5 V.

To measure the output power and to control the supply voltages, a 12-bit AD7896 U6 ADC from Analog Devices is used. One ADC channel is not enough, so a U5 switch is used at the input (it would be even better to use an 8-channel ADC, for example, of the AD7888 type). Data is read from the ADC in serial form. For this, the SDATA (serial data) and SCLK (clock signal) lines are used. The conversion process is started by software signal START. REF195 (U7) was used as a reference source and at the same time a voltage regulator for the ADC. Since the supply voltage of ±15 V is turned off in standby mode, all logic signals are connected to the ADC through resistors R9 - R11, which limit possible current surges when switching to standby mode and back.

Of the eight inputs of the switch, six are used: two for measuring power, four for monitoring supply voltages. The desired channel is selected using the address lines AX0, AX1, AX2.

Consider power measurement circuit left channel. The applied scheme ensures the multiplication of load current and voltage, so the load impedance is automatically taken into account and the readings always correspond to the real active power in the load. Through resistor dividers R49 - R54 located on the main board (Fig. 5), the voltage from the current sensors (emitter resistors of the output transistors) is fed to the U8A differential amplifier (Fig. 9), which emits a current signal. From the output of U8A through the trimmer resistor R17, the signal is fed to the input Y of the analog multiplier U9 of the K525PS2 type. The voltage signal is simply taken from the divider and fed to the X input of the analog multiplier. At the output of the multiplier, a R18C13 low-pass filter is installed, which extracts a signal proportional to the quasi-peak output power with an integration time of about 10 ms. This signal is fed to one of the switch inputs, then to the ADC. Diode VD1 protects the switch input from negative voltage.

In order to compensate for the initial zero offset of the multipliers, when the amplifier is turned on (when the load relay is not yet switched on and the output power is zero), the zero auto-calibration process occurs. The measured bias voltage during further operation is subtracted from the ADC readings.

The power in the left and right channels is measured separately, and the maximum value for the channels is displayed. Since the display must show both quasi-peak and average output power, and the displayed values ​​must be easy to read, the values ​​measured by the ADC are subject to software processing. The timing characteristics of a power meter are characterized by the integration time and the flyback time. For a quasi-peak power meter, the integration time is set by the hardware filtering circuit and is approximately 10 ms. The average power meter differs only in an increased integration time, which is implemented in software. When calculating the average power, a moving average of 256 points is used. The return time in both cases is set by software. For the convenience of reading the readings, this time should be relatively large. In this case, the reverse movement of the indicator is realized by subtracting 1/16 of the current power code once every 20 ms. In addition, during the indication, peak values ​​are held for 1.4 seconds. Since updating the indicator readings too often is badly perceived, the update occurs every 320 ms. In order not to miss the next peak and display it synchronously with the input signal, an extraordinary update of the readings occurs when a peak is detected.

As mentioned above, the PA uses a common with the preamplifier remote control, which works in the RC-5 standard. The remote control receiver type SFH-506 is located on the display board. From the output of the photodetector, the signal is fed to the input SER (INT1) of the microcontroller. Decoding of the RC-5 code is done by software. The system number used is 0AH, the STANDBY button is 0CH, the DISPLAY button is 21H, the MODE button is 20H. If necessary, these codes can be easily changed, since a conversion table is used, which can be found at the end of the source text of the microcontroller program.

On the display board(Fig. 10) two two-digit seven-segment indicators HG1 and HG2 type LTD6610E are installed. They are controlled by parallel registers U1 - U4. Dynamic indication is not used, as this may cause increased noise levels.

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Rice. 10. Schematic diagram of the indication board.

Register U5 is used to control the LEDs. A limiting resistor is connected in series with each segment and with each LED. The OC inputs of all registers are combined and connected to the PEN signal of the microcontroller. During the reset and initialization of the registers, this signal is in a logic high state. This prevents accidental ignition of the indication during transients.

The display board also has control buttons SB1 - SB6. They are connected to the data bus lines and to the RET return line. Diodes VD1 - VD6 prevent short circuit data lines when two or more buttons are pressed at the same time. When scanning the keyboard, the microcontroller uses port P0 as a simple output port, forming a running zero on its lines. The RET line is polled at the same time. Thus, the code of the pressed button is determined.

Next to the indicators, under a common protective glass, an integral remote control photodetector U6 is installed. The signal from the output of the photodetector through the XP6 connector is fed to the input of the microcontroller SER (INT1).

duty source(Fig. 11) provides 4 levels at the output: +5 V, +12 V and ±15 V. Levels of ±15 V are disabled in standby mode. The source uses a small toroidal transformer wound on a 50x20x25 mm core. The standby transformer has a large power margin, and the number of turns per volt is chosen more than the calculated one. Thanks to these measures, the transformer practically does not heat up, which increases its reliability (after all, it must work continuously throughout the entire life of the amplifier). Winding data and wire diameter are indicated in the diagram. Voltage stabilizers have no features. Stabilizer circuits U1 and U2 are mounted on a small common radiator. To turn off the levels of ±15 V, switches on transistors VT1 - VT4 are used, which are controlled by the PEN signal coming from the processor board.

Rice. 11. Schematic diagram of the standby power supply board.

In addition to voltage stabilizers, switches on transistors VT5 - VT12 are installed on the standby power supply board to control the relay and fan. Since the microcontrollers of the MCS-51 family, during the action of the “Reset” signal, the ports are in a logic high state, all actuators must turn on at a low level. Otherwise, there will be false positives at the moment of power-up or if the watchdog timer is triggered. For this reason, single npn transistors with OE or ULN2003 driver chips and the like cannot be used as keys.

Relays, fuses and limiting resistors are located on relay board(Fig. 12). Connection of all network wires is made through screw terminal blocks. Each main transformer, duty transformer and external socket block have separate fuses. For safety reasons, external sockets are switched off by two groups of relay contacts K1, which break both wires. The main transformers have a tap from the middle of the primary winding. This tap can be used to supply 110 V to power other components of the complex. Devices that meet the American standard are somewhat cheaper than multi-system ones, so they are sometimes found on our territory. There are points on the relay board from which 110 V can be removed, but this voltage is not used in the basic version.

Rice. 12. Schematic diagram of the relay board.

Block connection diagram on amplifier chassis shown in fig. 13. Bridge rectifiers assembled on diodes VD5 - VD12 of the KD2997A type are connected to the secondary windings of the main transformers T1 and T2. Filter capacitors with a total capacity of more than 100,000 uF are connected to the output of the rectifiers. This high capacitance is needed to achieve low ripple and improve the amplifier's ability to reproduce pulsed signals. From the filter capacitors, the supply voltage of ±45 V is supplied to the main amplifier boards. Additionally, there are low-power rectifiers assembled on diodes VD1 - VD4, the output voltage of which is filtered with a relatively small time constant by capacitors C1 and C2. Through resistors R1 and R2, the output voltage of these auxiliary rectifiers is fed to the protection circuits that are assembled on the main amplifier boards. When several half-cycles of the mains voltage fail, the output voltage of the auxiliary rectifiers drops, which is detected by the protection circuits, and the load relays are disconnected. At this time, the output voltage of the main rectifiers is still quite high due to large capacitors, so the transient process in the amplifier does not start when the load is connected.

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Rice. 13. Connection diagram of amplifier blocks.

For power amplifier design and layout no less important than circuitry. The main problem is that the output transistors require efficient heat dissipation. With the natural method of cooling, this results in massive radiators, which become almost the main structural elements. The common layout, when the rear wall serves as a radiator at the same time, is not suitable, since then there is no room at the back for installing the necessary terminals and connectors. Therefore, in the described UM, a layout with a lateral location of radiators was chosen (Fig. 14):

Rice. 14. General layout of the amplifier.

The radiators are slightly raised (this is clearly seen in Fig. 4), which ensures their better cooling. The main amplifier boards are fixed parallel to the radiators. This minimizes the length of wires between the board and the output transistors. Another dimensional elements of the amplifier are network transformers. In this case, two toroidal transformers are used, which are installed on top of each other in a common cylindrical screen. This screen occupies a significant part of the internal volume of the amplifier case. The main rectifiers are mounted on a common radiator, which is located vertically behind the screen of the transformers. The filter capacitors are located at the bottom of the amplifier chassis and are covered by a tray. The relay board is also located there. The standby power supply is mounted on a special bracket near the rear panel. The processor and display boards are located in the thickness of the front panel, which has a box-shaped section.

When developing the design of the amplifier, much attention was paid to the manufacturability of the design and ease of access to any node. More details about the layout of the amplifier can be found in Fig. 15 and 18:

Rice. 15. Arrangement of assembled amplifier nodes.

The basis of the amplifier case is aluminum alloy chassis D16T 4mm thick (4 in Fig. 18). Attached to the chassis radiators(1 in Fig. 18) which are milled from an aluminum plate or casting. The required area of ​​the radiators strongly depends on the operating conditions of the amplifier, but it should not be less than 2000 cm 2. To facilitate access to the amplifier boards, the heatsinks are attached to the chassis with hinges (10 in Fig. 18), which allows the heatsinks to be tilted out. In order not to interfere with the wires of the input and output connectors, rear panel divided into three parts (Fig. 4). The middle part is fixed with a bracket on the chassis, and the two side parts are fixed on the radiators. Connectors are installed on the sides of the panel, which fold out along with the heatsinks. Thus, the radiator assembly is a monophonic PA, which is connected only by power wires and a flat control cable. On fig. 18, for clarity, the radiators are only partially folded back, and the rear panel is not disassembled.

Main amplifier boards are also attached to the heatsinks with hinges (12 in Fig. 18), which allows them to be folded back to gain access to the soldering side. The axis of rotation of the board runs along the line of holes for connecting the wires of the output transistors. This made it possible to practically not increase the length of these wires while at the same time being able to tilt the board. The upper mounting points of the boards are ordinary threaded racks with a height of 15mm. The layout of the single-sided main boards of the left and right channels is done mirror(Fig. 16), which made it possible to optimize the connections. Naturally, the mirroring of the topology is not complete, since elements are used that cannot simply be mirrored (microcircuits and relays). The figure gives a rough idea of ​​the topology of the boards, the topology of all the boards is available in the archive (see the Download section) as files in the PCAD 4.5 format.

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Rice. 16. Wiring of the main boards of the amplifier.

Each radiator 1 (Fig. 17) has a smooth surface 2, which is processed after blackening. Nine transistors 4 are installed on it through ceramic gaskets 2.

Rice. 17. Design of radiators:

Studies have shown that mica, and even more so modern elastic gaskets, do not have sufficient thermal conductivity. The best material for insulating gaskets is BeO-based ceramic. However, for transistors in plastic cases, such gaskets are almost never found. Pretty good results were obtained by fabricating gaskets from hybrid microcircuit substrates. This is a pink ceramic (unfortunately, the material is not exactly known, most likely something based on Al 2 O 3). To compare the thermal conductivity of different gaskets, a stand was assembled, in which two identical transistors were fixed on the radiator in the TO-220 package: one directly, the other through the test gasket. The base current for both transistors was the same. The transistor on the pad dissipated about 20W of power, while the other transistor did not dissipate power (no voltage was applied to the collector). The difference between the B-E drops for two transistors was measured, and the difference in junction temperatures was calculated from this difference. Thermal paste was used for all gaskets, without it the results were worse and unstable. The comparison results are presented in the table:

The output transistors are pressed with pads 5, the rest of the transistors are fastened with screws. This is not very convenient, since drilling of ceramic gaskets is required, which can only be done with the help of diamond drills, and even then with great difficulty.

A thermometer 9 is installed next to the transistors. As experience has shown, when attaching DS1820 thermometers, high pressure cannot be exerted on their case, otherwise the readings are distorted, and very significantly (it is better to glue the thermometers with a glue that has high thermal conductivity).

Board 6 is fixed under the transistors on the radiator. There are no conductors on the reverse side of this board, so it can be mounted directly on the surface of the radiator. The outputs of all transistors are soldered to the pads on the top side of the board. The connections of the board to the main board are made with short wires, which are soldered into hollow rivets 7. To prevent the rivets from shorting to the radiator, a recess 8 is made in it.

Basic toroidal transformers(7 in Fig. 18) are installed on top of each other through elastic pads. To reduce interference from transformers to other equipment (cassette deck, for example), it is recommended to place the transformers in a screen made of annealed steel with a thickness of at least 1.5 mm. The screen is a steel cylinder and two covers, pulled together with a pin. To avoid the appearance of a short-circuited coil, the top cover has a dielectric sleeve. However, if it is supposed to operate the PA at high average power, then ventilation holes should be provided in the screen or the screen should be completely abandoned. It would seem that in order to mutually compensate for the leakage fields of transformers, it is enough to simply turn on their primary windings in antiphase. But in practice, this measure is very ineffective. The stray field of a toroidal transformer, with its apparent axial symmetry, has a very complex spatial distribution. Therefore, the polarity reversal of one of the primary windings leads to a weakening of the stray field at one point in space, but to an increase in another. In addition, the configuration of the stray field depends significantly on the load of the transformer.

Rice. 18. The main components of the amplifier:

Very stringent requirements are imposed on the UM power transformer. This is due to the fact that it is loaded on a rectifier with very large filter capacitors. This leads to the fact that the current consumed from the secondary winding of the transformer is of a pulsed nature, and the value of the current in the pulse is many times higher than the average current consumed. To keep transformer losses low, the windings must have very low resistance. In other words, the transformer must be designed for significantly more power than is consumed on average from it. In the described amplifier, two toroidal transformers are used, each of which is wound on a core 110x60x40 mm from E-380 steel tape. The primary windings contain 2x440

UMZCH VV with microcontroller control system
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UMZCH VVS-2011 Ultimate version

Amplifier Specifications:

Big Power: 150W/8ohm
High linearity: 0.0002 - 0.0003% (at 20 kHz 100 W / 4 ohms)

Full set of service nodes:

Maintain zero DC voltage
AC wire resistance compensator
current protection
Output DC voltage protection
Smooth start

Wiring diagram

The layout of printed circuit boards was carried out by a participant in many popular projects LepekhinV (Vladimir Lepekhin). It worked out very well).

VVS-2011 Amplifier Board

Starter protection device

VVS-2011 Amplifier AC Protection Board

The ULF VVS-2011 amplifier board was designed for tunnel blowing (parallel to the radiator). The installation of transistors UN (voltage amplifier) ​​and VK (output stage) is somewhat difficult, because. installation / dismantling has to be done with a screwdriver through holes in the PCB with a diameter of about 6 mm. When access is open, the projection of transistors does not fall under the PP, it is much more convenient. I had to tweak the board a bit.

amplifier board

Wiring diagram of the VVS-2011 amplifier

In the new software, one point was not taken into account - this is the convenience of setting up protection on the amplifier board

C25 \u003d 0.1 nF, R42 * \u003d 820 Ohm and R41 \u003d 1 kOhm. All smd elements are located on the soldering side, which is not very convenient when setting up, because. it will be necessary to unscrew and fasten the bolts of the PCB on the racks and transistors to the radiators several times.

Sentence: R42 * 820 Ohm consists of two smd resistors located in parallel, from here the proposal: we solder one smd resistor immediately, solder the other output resistor with a canopy to VT10, one lead to the base, the other to the emitter, we select to the appropriate one. We picked it up, we change the output to smd, for clarity.

Viktor Zhukovsky, Krasnoarmeysk, Donetsk region

UMZCH BB-2010 is a new development from the well-known line of amplifiers UMZCH BB (high fidelity) [1; 2; 5]. A number of technical solutions used were influenced by the work of Ageev SI. .

The amplifier provides Kr of the order of 0.001% at a frequency of 20 kHz with Рout = 150 W at a load of 8 ohms, a small signal bandwidth at a level of -3 dB - 0 Hz ... 800 kHz, an output voltage slew rate of -100 V / μs, signal-to-noise ratio and signal/background -120 dB.

Due to the use of an op amp operating in a light mode, as well as the use of only stages with OK and OB in the voltage amplifier, covered by deep local OOS, UMZCH BB is highly linear even before the general OOS is covered. In the very first high-fidelity amplifier back in 1985, solutions were used that until then were used only in measuring technology: DC modes are supported by a separate service node, to reduce the level of interface distortions, the transient resistance of the AC switching relay contact group is covered by a common negative feedback, and a special node effectively compensates for the influence of the resistance of the AC cables on these distortions. The tradition has been preserved in UMZCH BB-2010, however, the general environmental protection also covers the resistance of the output low-pass filter.

In the vast majority of designs of other UMZCH, both professional and amateur, many of these solutions are still missing. At the same time, the high technical characteristics and audiophile advantages of the UMZCH BB are achieved by simple circuit solutions and a minimum of active elements. In fact, this is a relatively simple amplifier: one channel can be assembled slowly in a couple of days, and the setting consists only in setting the required quiescent current of the output transistors. Especially for beginner radio amateurs, a method has been developed for node-by-node, cascade-based performance testing and adjustment, using which you can guarantee to localize the places of possible errors and prevent their possible consequences even before the UMZCH is fully assembled. For all possible questions about this or similar amplifiers, there are detailed explanations, both on paper and on the Internet.

At the input of the amplifier, an R1C1 high-pass filter with a cutoff frequency of 1.6 Hz is provided, Fig. 1. But the efficiency of the mode stabilization device allows the amplifier to work with an input signal containing up to 400 mV of DC voltage. Therefore, C1 is excluded, which realizes the age-old audiophile dream of a path without capacitors © and significantly improves the sound of the amplifier.

The capacitance of the capacitor C2 of the input low-pass filter R2C2 is chosen so that the cutoff frequency of the input low-pass filter, taking into account the output resistance of the preamplifier 500 Ohm -1 kOhm, is in the range from 120 to 200 kHz. The frequency correction circuit R3R5C3 is placed at the input of the op-amp DA1, which limits the band of processed harmonics and interference coming through the CUS circuit from the output side of the UMZCH to a band of 215 kHz at a level of -3 dB and increases the stability of the amplifier. This circuit makes it possible to reduce the difference signal above the cutoff frequency of the circuit and thus eliminate unnecessary overloading of the voltage amplifier by high-frequency interference, noise and harmonics, eliminating the possibility of dynamic intermodulation distortion (TIM; DIM).

Next, the signal is fed to the input of a low-noise operational amplifier with field-effect transistors at the input DA1. Many "claims" against the UMZCH BB are made by opponents regarding the use of an op-amp at the input, which allegedly degrades the sound quality and "steals the virtual depth" of the sound. In this regard, it is necessary to pay attention to some quite obvious features of the operation of the OS in the UMZCH VV.

Operational amplifiers of preamplifiers, post-DAC op-amps are forced to develop several volts of output voltage. Since the gain of the op amps is low, ranging from 500 to 2,000 times at 20 kHz, this indicates that they operate with a relatively large difference signal voltage - from several hundred microvolts at low frequencies to several millivolts at 20 kHz, and a high probability of introducing intermodulation distortion by the op amp input stage. The output voltage of these op amps is equal to the output voltage of the last voltage amplification stage, usually made according to the OE scheme. An output voltage of several volts indicates the operation of this cascade with rather large input and output voltages, and as a result, it introduces distortions into the amplified signal. The op-amp is loaded with the resistance of the OOS circuit and the load connected in parallel, sometimes amounting to several kilo-ohms, which requires up to several milliamps from the output follower of the output current amplifier. Therefore, changes in the current of the output follower of the IC, the output stages of which consume a current of no more than 2 mA, are quite significant, which also indicates that they introduce distortions into the amplified signal. We see that the input stage, the voltage amplification stage and the output stage of the op-amp can introduce distortions.

But the high-fidelity amplifier circuitry, due to the high gain and input resistance of the transistor part of the voltage amplifier, provides very gentle operating conditions for the op-amp DA1. Judge for yourself. Even in the UMZCH, which has developed a rated output voltage of 50 V, the input differential stage of the op amp operates with differential voltage signals from 12 μV at frequencies of 500 Hz to 500 μV at a frequency of 20 kHz. The ratio of the high input overload capacity of the differential stage, made on field-effect transistors, and the meager voltage of the difference signal provides a high linearity of signal amplification. The output voltage of the op-amp does not exceed 300 mV. which indicates a low input voltage of the voltage amplification stage with a common emitter from the operational amplifier - up to 60 μV - and the linear mode of its operation. The output stage of the op-amp gives to the load about 100 kOhm from the side of the VT2 base an alternating current of not more than 3 μA. Consequently, the output stage of the op-amp also operates in an extremely lightweight mode, almost at idle. On a real musical signal, voltages and currents are, most of the time, an order of magnitude less than the given values.

From a comparison of the voltages of the difference and output signals, as well as the load current, it can be seen that, in general, the operational amplifier in the UMZCH BB operates hundreds of times easier, and, therefore, in a linear mode than the op-amp mode of preamplifiers and post-DAC op-amps of CD players that serve as sources signal for UMZCH with any depth of environmental protection, as well as without it at all. Consequently, the same op amp will introduce much less distortion as part of the UMZCH BB than in a single inclusion.

Occasionally there is an opinion that the distortions introduced by the cascade are ambiguously dependent on the voltage of the input signal. This is mistake. The dependence of the manifestation of the nonlinearity of the cascade on the voltage of the input signal may obey one law or another, but it is always unambiguous: an increase in this voltage never leads to a decrease in the introduced distortions, but only to an increase.

It is known that the level of distortion products attributable to a given frequency decreases in proportion to the depth of negative feedback for this frequency. The idle speed gain, up to the coverage of the feedback amplifier, at low frequencies cannot be measured due to the smallness of the input signal. According to calculations, the idle amplification developed up to the NOS coverage makes it possible to achieve an OOS depth of 104 dB at frequencies up to 500 Hz. Measurements for frequencies starting from 10 kHz show that the depth of feedback at a frequency of 10 kHz reaches 80 dB, at a frequency of 20 kHz - 72 dB, at a frequency of 50 kHz - 62 dB and 40 dB - at a frequency of 200 kHz. Figure 2 shows the amplitude-frequency characteristics of UMZCH BB-2010 and, for comparison, UMZCH similar in complexity to Leonid Zuev.

High gain before coverage of the OOS is the main feature of the circuit design of VV amplifiers. Since the goal of all circuitry tricks is to achieve high linearity and high gain for maintaining deep feedback in the widest possible frequency band, this means that circuitry methods for improving amplifier parameters are exhausted by such structures. Further reduction of distortion can only be ensured by constructive measures aimed at reducing the pickup of harmonics of the output stage on the input circuits, especially on the inverting input circuit, the gain from which is maximum.

Another feature of the UMZCH BB circuitry is the current control of the output stage of the voltage amplifier. The input op-amp controls the voltage-to-current conversion stage, performed with OK and OB, and the received current is subtracted from the quiescent current of the stage, performed according to the OB circuit.

The use of a linearizing resistor R17 with a resistance of 1 kOhm in the differential stage VT1, VT2 on transistors of different structures with serial power increases the linearity of the conversion of the output voltage of the op-amp DA1 to the collector current VT2 by creating a local OOS with a depth of 40 dB. This can be seen from a comparison of the sum of the intrinsic resistances of the emitters VT1, VT2 - approximately 5 ohms each - with the resistance R17, or the sum of the thermal voltages VT1, VT2 - about 50 mV - with a voltage drop across the resistance R17, which is 5.2 - 5.6 V .

Amplifiers built according to the considered circuitry have a sharp, 40 dB per decade of frequency, gain decay above a frequency of 13 ... 16 kHz. The error signal, which is a distortion product, at frequencies above 20 kHz is two to three orders of magnitude smaller than the useful audio signal. This makes it possible to convert the linearity of the differential stage VT1, VT2, which is excessive at these frequencies, into an increase in the gain of the transistor part of the UN. Due to slight changes in the current of the differential stage VT1, VT2, when weak signals are amplified, its linearity does not deteriorate significantly with a decrease in the depth of the local OOS, but the operation of the op-amp DA1, on the operating mode of which the linearity of the entire amplifier depends on the operating mode of which at these frequencies, the gain margin will facilitate, since all voltages, The distortions that determine the distortions introduced by the operational amplifier, starting from the difference signal to the output signal, decrease in proportion to the gain in gain at a given frequency.

The phase advance correction circuits R18C13 and R19C16 were optimized in the simulator in order to reduce the difference voltage of the op-amp to frequencies of several megahertz. It was possible to increase the gain of UMZCH BB-2010 compared to UMZCH BB-2008 at frequencies of the order of several hundred kilohertz. Gain gain was 4 dB at 200 kHz, 6 dB at 300 kHz, 8.6 dB at 500 kHz, 10.5 dB at 800 kHz, 11 dB at 1 MHz, and 10 to 12 dB at frequencies above 2 MHz. This can be seen from the simulation results, Fig. 3, where the lower curve refers to the frequency response of the UMZCH BB-2008 lead correction circuit, and the upper one to UMZCH BB-2010.

VD7 protects the emitter junction VT1 from reverse voltage arising from the flow of recharging currents C13, C16 in the voltage limiting mode of the UMZCH output signal and the resulting limit voltages with a high rate of change at the output of the op-amp DA1.

The output stage of the voltage amplifier is made on a transistor VT3, connected according to a common base circuit, which excludes the penetration of a signal from the output circuits of the stage into the input circuits and increases its stability. The cascade with OB, loaded on the current generator on the transistor VT5 and the input impedance of the output stage, develops a high stable gain - up to 13,000 ... 15,000 times. The choice of the resistance of the resistor R24 ​​half the resistance of the resistor R26 guarantees the equality of the quiescent currents VT1, VT2 and VT3, VT5. R24, R26 provide local OOS that reduce the effect of the Earley effect - the change in p21e depending on the collector voltage and increase the initial linearity of the amplifier by 40 dB and 46 dB, respectively. The supply of the UN with a separate voltage, modulo 15 V higher than the voltage of the output stages, makes it possible to eliminate the effect of quasi-saturation of transistors VT3, VT5, which manifests itself in a decrease in n21e when the collector-base voltage drops below 7 V.

The three-stage output follower is assembled on bipolar transistors and does not require any special comments. Don't try to fight entropy © by saving on the quiescent current of the output transistors. It should not be less than 250 mA; in the author's version - 320 mA.

Prior to the operation of the relay for switching on AC K1, the amplifier is covered by OOS1, implemented by turning on the divider R6R4. The accuracy of maintaining the resistance R6 and the consistency of these resistances in different channels is not essential, but to maintain the stability of the amplifier it is important that the resistance R6 is not much lower than the sum of the resistances R8 and R70. By actuating relay K1, the OOS1 is turned off and the OOS2 circuit, formed by R8R70C44 and R4, comes into operation, and covers the contact group K1.1, where R70C44 excludes the output low-pass filter R71L1 R72C47 from the OOS circuit at frequencies above 33 kHz. The frequency-dependent OOS R7C10 generates a decline in the frequency response of the UMZCH to the output low-pass filter at a frequency of 800 kHz at a level of -3 dB and provides a margin in depth of the OOS above this frequency. The frequency response decay at the AC terminals above the frequency of 280 kHz at a level of -3 dB is provided by the combined action of the R7C10 and the output low-pass filter R71L1 -R72C47.

The resonant properties of loudspeakers lead to the emission of damped sound vibrations by the diffuser, overtones after impulse action and the generation of its own voltage when the turns of the loudspeaker coil cross the magnetic field lines in the gap of the magnetic system. The damping coefficient shows how large the amplitude of the diffuser oscillations is and how quickly they decay when the AC is loaded as a generator on the impedance from the UMZCH. This coefficient is equal to the ratio of the AC resistance to the sum of the output resistance of the UMZCH, the transient resistance of the contact group of the AC switching relay, the resistance of the inductor coil of the output LPF usually wound with a wire of insufficient diameter, the transient resistance of the AC cable clamps and the resistance of the AC cables themselves.

In addition, the impedance of loudspeakers is non-linear. The flow of distorted currents through the wires of AC cables creates a voltage drop with a high degree of non-linear distortion, which is also subtracted from the undistorted output voltage of the amplifier. Therefore, the signal at the AC terminals is much more distorted than at the UMZCH output. These are the so-called interface distortions.

To reduce these distortions, compensation of all components of the total output impedance of the amplifier was applied. The own output resistance of the UMZCH, together with the contact resistance of the relay contacts and the resistance of the wire of the inductor of the output low-pass filter, is reduced by the action of a deep general OOS taken from the right output of L1. In addition, by connecting the right output of R70 to the “hot” AC terminal, you can easily compensate for the transient resistance of the AC cable clamp and the resistance of one of the AC wires, without fear of generating UMZCH due to phase shifts in the wires covered by the OOS.

The AC wire resistance compensation unit is made in the form of an inverting amplifier with Ky = -2 on the DA2, R10, C4, R11 and R9 op-amps. The input voltage for this amplifier is the voltage drop on the "cold" ("earth") wire of the speaker. Since its resistance is equal to the resistance of the "hot" wire of the AC cable, to compensate for the resistance of both wires, it is enough to double the voltage on the "cold" wire, invert it and through the resistor R9 with a resistance equal to the sum of the resistances R8 and R70 of the OOS circuit, apply to the inverting input of the op-amp DA1 . Then the output voltage of the UMZCH will increase by the sum of the voltage drops on the AC wires, which is equivalent to eliminating the influence of their resistance on the damping coefficient and the level of interface distortion at the AC terminals. Compensation for the drop in the resistance of the AC wires of the non-linear component of the back-EMF of loudspeakers is especially needed at the lower frequencies of the audio range. The signal voltage at the tweeter is limited by a resistor and capacitor connected in series with it. Their complex resistance is much greater than the resistance of the wires of the AC cable, so the compensation of this resistance at the RF is meaningless. Based on this, the integrating circuit R11C4 limits the operating frequency band of the compensator to 22 kHz.

Of particular note: the resistance of the "hot" wire of the AC cable can be compensated by covering it with a common OOS by connecting the right terminal of R70 with a special wire to the "hot" AC terminal. In this case, only the resistance of the "cold" AC wire will need to be compensated, and the gain of the wire resistance compensator must be reduced to the value Ku \u003d -1 by choosing the resistance of the resistor R10 equal to the resistance of the resistor R11.

The current protection unit prevents damage to the output transistors during short circuits in the load. Resistors R53 - R56 and R57 - R60 serve as a current sensor, which is quite enough. The amplifier output current flowing through these resistors creates a voltage drop that is applied to the divider R41R42. A voltage with a value greater than the threshold opens the transistor VT10, and its collector current opens the VT8 trigger cell VT8VT9. This cell goes into a steady state with open transistors and shunts the HL1VD8 circuit, reducing the current through the zener diode to zero and locking VT3. Discharging C21 with a small base current VT3 can take a few milliseconds. After the trigger cell is activated, the voltage on the lower plate of C23, charged by the voltage on the HL1 LED to 1.6 V, rises from the level of -7.2 V from the positive power rail of the UN to the level of -1.2 V 1, the voltage on the upper plate of this capacitor also rises to 5 V. C21 is quickly discharged through the resistor R30 to C23, the transistor VT3 is locked. Meanwhile, VT6 opens and through R33, R36 opens VT7. VT7 shunts the zener diode VD9, discharges capacitor C22 through R31 and turns off transistor VT5. Not receiving a bias voltage, the output stage transistors are also locked.

Restoring the initial state of the trigger and turning on the UMZCH is done by pressing the button SA1 "Reset protection". C27 is charged by the VT9 collector current and shunts the VT8 base circuit, locking the trigger cell. If by this time the emergency has been eliminated and VT10 is locked, the cell goes into a state with stably closed transistors. VT6, VT7 are closed, a reference voltage is applied to the bases VT3, VT5 and the amplifier enters the operating mode. If the short circuit in the UMZCH load continues, the protection is activated again, even if the capacitor C27 is connected to SA1. The protection works so effectively that during the adjustment of the correction, the amplifier was de-energized several times for small soldering ... by touching the non-inverting input. The resulting self-excitation led to an increase in the current of the output transistors, and the protection turned off the amplifier. Although this crude method should not be offered as a rule, but due to current protection, it did not harm the output transistors.

The work of the compensator for the resistance of AC cables.

The efficiency of the UMZCH BB-2008 compensator was tested by the old audiophile method, by ear, by switching the compensator input between the compensating wire and the common wire of the amplifier. The improvement in sound was clearly noticeable, and the future owner was eager to get an amplifier, so no measurements of the effect of the compensator were carried out. The advantages of the cable-cutter scheme were so obvious that the compensator + integrator configuration was adopted as the standard assembly for installation in all developed amplifiers.

It's amazing how much unnecessary debate about the usefulness / uselessness of cable resistance compensation has flared up on the Internet. As usual, those to whom the extremely simple cable-cleaning scheme seemed complicated and incomprehensible, the costs for it - exorbitant, and the installation - time-consuming ©, especially insisted on listening to a non-linear signal. There were even suggestions that, since so much money is being spent on the amplifier itself, it’s a sin to save on the sacred, but you need to go the best, glamorous way that all civilized mankind goes and ... buy normal, human © super-expensive cables made of precious metals. To my great surprise, the statements of highly respected experts about the uselessness of the compensation unit at home, including those specialists who successfully use this unit in their amplifiers, added fuel to the fire. It is very unfortunate that many fellow radio amateurs were distrustful of reports about improving the sound quality at low and medium frequencies with the inclusion of a compensator, avoided this simple way to improve the operation of the UMZCH with all their might, than robbed themselves.

Little research has been done to document the truth. A number of frequencies were supplied from the GZ-118 generator to the UMZCH BB-2010 in the region of the AC resonant frequency, the voltage was controlled by an S1-117 oscilloscope, and Kr at the AC terminals was measured by INI C6-8, Fig. 4. Resistor R1 is installed to avoid interference at the input of the compensator when switching it between the control and common wires. The experiment used common and publicly available AC cables with a length of 3 m and a core cross section of 6 square meters. mm, as well as the GIGA FS Il speaker system with a frequency range of 25 -22.000 Hz, a nominal impedance of 8 ohms and a rated power of 90 W from Acoustic Kingdom.

Unfortunately, the circuitry of the harmonic signal amplifiers from the C6-8 composition provides for the use of high-capacity oxide capacitors in the environmental protection circuits. This causes the low-frequency noise of these capacitors to affect the resolution of the device at low frequencies, as a result of which its resolution at low frequencies deteriorates. When measuring Kr of a signal with a frequency of 25 Hz from GZ-118 directly from C6-8, the instrument readings dance around a value of 0.02%. It is not possible to get around this limitation using the GZ-118 generator notch filter in the case of measuring the compensator efficiency, because a number of discrete values ​​of the tuning frequencies of the 2T filter are limited at low frequencies by the values ​​of 20.60, 120, 200 Hz and do not allow measuring Kr at the frequencies of interest to us. Therefore, reluctantly, the level of 0.02% was taken as zero, the reference.

At a frequency of 20 Hz with a voltage at the AC terminals of 3 Vpp, which corresponds to an output power of 0.56 W into an 8 ohm load, Kr was 0.02% with the compensator on and 0.06% after it was turned off. At a voltage of 10 V amps, which corresponds to an output power of 6.25 W, the Kr value is 0.02% and 0.08%, respectively, at a voltage of 20 V amps and a power of 25 W - 0.016% and 0.11%, and at a voltage of 30 In the amplitude and power of 56 W - 0.02% and 0.13%.

Knowing the relaxed attitude of manufacturers of imported equipment to the values ​​​​of inscriptions regarding power, and also remembering the miraculous, after the adoption of Western standards, the transformation of the 35AC-1 speaker system with a subwoofer power of 30 W into S-90, long-term power of more than 56 W was not supplied to AC.

At a frequency of 25 Hz at a power of 25 W, Kr was 0.02% and 0.12% with the compensation unit on / off, and at a power of 56 W - 0.02% and 0.15%.

At the same time, the necessity and effectiveness of covering the output LPF of the general OOS was checked. At a frequency of 25 Hz at a power of 56 W and connected in series to one of the wires of the AC cable of the output RL-RC low-pass filter, similar to that installed in the superlinear UMZCH, Kr with the compensator turned off reaches 0.18%. At a frequency of 30 Hz at a power of 56 W Kr 0.02% and 0.06% with the compensation unit on / off. At a frequency of 35 Hz at a power of 56 W, Kr is 0.02% and 0.04% with the compensation unit on / off. At frequencies of 40 and 90 Hz at a power of 56 W, Kr is 0.02% and 0.04% with the compensation unit on / off, and at a frequency of 60 Hz - 0.02% and 0.06%.

The conclusions are obvious. There is a presence of non-linear distortion of the signal at the AC terminals. The deterioration of the linearity of the signal at the AC terminals is clearly recorded with its inclusion through an uncompensated, uncovered OOS resistance of a low-pass filter containing 70 cm of a relatively thin wire. The dependence of the level of distortion on the power supplied to the AC suggests that it depends on the ratio of the signal power and the nominal power of the AC woofers. Distortions are most pronounced at frequencies near the resonant one. The back EMF generated by the speakers in response to the impact of an audio signal is shunted by the sum of the output resistance of the UMZCH and the resistance of the wires of the AC cable, so the level of distortion at the AC terminals directly depends on the resistance of these wires and the output impedance of the amplifier.

The cone of a poorly damped woofer itself emits overtones, and in addition, this loudspeaker generates a wide tail of harmonics and intermodulation distortion products that a midrange loudspeaker reproduces. This explains the deterioration of the sound at medium frequencies.

Despite the assumption of a zero Kr level of 0.02% due to the imperfection of the IRI, the effect of the cable resistance compensator on signal distortion at the AC terminals is clearly and unambiguously noted. It can be stated that the conclusions made after listening to the operation of the compensation unit on a musical signal and the results of instrumental measurements are in full agreement.

The improvement that is clearly audible when the cable cleaner is turned on can be explained by the fact that with the disappearance of distortion on the AC terminals, the midrange loudspeaker stops reproducing all this dirt. Apparently, therefore, by reducing or eliminating the reproduction of distortions by a mid-frequency loudspeaker, a two-cable AC connection circuit, the so-called. "biwiring", when the LF and MF-HF links are connected by different cables, has an advantage in sound compared to a single-cable circuit. However, since in a two-cable circuit the distorted signal at the terminals of the LF section of the AC does not disappear anywhere, this circuit loses to the option with a compensator in terms of the damping coefficient of the free vibrations of the cone of the low-frequency loudspeaker.

You can't deceive physics, and for a decent sound it is not enough to get brilliant performance at the output of the amplifier with an active load, but it is also necessary not to lose linearity after the signal is delivered to the speaker terminals. As part of a good amplifier, a compensator made according to one scheme or another is absolutely necessary.

Integrator.

The effectiveness and possibility of reducing the error of the DA3 integrator was also tested. In UMZCH BB with op-amp TL071, the output DC voltage is in the range of 6 ... 9 mV, and it was not possible to reduce this voltage by including an additional resistor in the non-inverting input circuit.

The effect of low-frequency noise characteristic of a DC-input op-amp, due to the coverage of deep feedback through the frequency-dependent circuit R16R13C5C6, manifests itself in the form of an instability of the output voltage of a few millivolts, or -60 dB relative to the output voltage at rated output power, at frequencies below 1 Hz , not reproducible speakers.

On the Internet, it was mentioned about the low resistance of the protective diodes VD1 ... VD4, which allegedly introduces an error into the operation of the integrator due to the formation of a divider (R16 + R13) / R VD2 | VD4 . . To check the reverse resistance of protective diodes, a circuit was assembled in Fig. 6. Here, the op-amp DA1, connected according to the inverting amplifier circuit, is covered by the OOS through R2, its output voltage is proportional to the current in the circuit of the tested diode VD2 and the protective resistor R2 with a coefficient of 1 mV / nA, and the resistance of the circuit R2VD2 - with a coefficient of 1 mV / 15 GΩ. To eliminate the influence of the additive errors of the op-amp - bias voltage and input current on the results of measuring the diode leakage current, it is necessary to calculate only the difference between the intrinsic voltage at the output of the op-amp, measured without the diode under test, and the voltage at the output of the op-amp after its installation. In practice, a difference in the output voltages of the op-amp of several millivolts gives the value of the reverse resistance of the diode of the order of ten to fifteen gigaohms at a reverse voltage of 15 V. It is obvious that the leakage current will not increase with a decrease in the voltage across the diode to a level of several millivolts, which is characteristic of the difference voltage of the op-amp of the integrator and compensator .

But the photoelectric effect inherent in diodes placed in a glass case really leads to a significant change in the output voltage of the UMZCH. When illuminated with an incandescent lamp of 60 W from a distance of 20 cm, the constant voltage at the output of the UMZCH increased to 20 ... 3O mV. Although it is unlikely that a similar level of illumination can be observed inside the amplifier case, a drop of paint applied to these diodes eliminated the dependence of the UMZCH modes on illumination. According to the simulation results, no drop in the frequency response of the UMZCH is observed even at a frequency of 1 millihertz. But the time constant R16R13C5C6 should not be reduced. The phases of the alternating voltages at the outputs of the integrator and the compensator are opposite, and with a decrease in the capacitance of the capacitors or the resistance of the resistors of the integrator, an increase in its output voltage can worsen the compensation of the resistance of the AC cables.

Amplifier sound comparison. The sound of the assembled amplifier was compared with the sound of several foreign industrial amplifiers. The source was a Cambridge Audio CD player, the Radiotekhnika UP-001 pre-amplifier was used to build up and adjust the sound level of the terminal UMZCH, the Sugden A21a and NAD C352 used regular adjustment controls.

The first to check was the legendary, outrageous and damn expensive English UMZCH "Sugden A21a", operating in class A with an output power of 25 watts. Remarkably, in the accompanying documentation for VCL, the British considered it good not to indicate the level of non-linear distortion. Say, it's not about distortions, but about spirituality. "Sugden A21a>" lost to UMZCH BB-2010 with comparable power both in terms of level and clarity, confidence, nobility of sound at low frequencies. This is not surprising, given the peculiarities of its circuitry: just a two-stage quasi-symmetric output follower on transistors of the same structure, assembled according to the circuitry of the 70s of the last century with a relatively high output impedance and an electrolytic capacitor switched on at the output that further increases the total output resistance - this is the last the solution itself degrades the sound of any amplifiers at low and medium frequencies. At medium and high frequencies, UMZCH BB showed higher detail, transparency and excellent stage elaboration, when singers and instruments could be clearly localized in sound. By the way, speaking of the correlation of objective measurement data and subjective impressions of the sound: in one of the magazine articles of Sugden's competitors, its Kr was determined at the level of 0.03% at a frequency of 10 kHz.

The next was also the English amplifier NAD С352. The general impression was the same: the pronounced "bucket" sound of the Englishman at low frequencies did not leave him any chances, while the work of the UMZCH BB was recognized as impeccable. Unlike NADa, whose sound was associated with thick bushes, wool, cotton wool, the sound of BB-2010 at medium and high frequencies made it possible to clearly distinguish the voices of performers in the general choir and instruments in the orchestra. In the work of NAD C352, the effect of better audibility of a more vociferous performer, a louder instrument, was clearly expressed. As the owner of the amplifier himself put it, in the sound of the UMZCH BB, the vocalists did not “shout out” to each other, and the violin did not fight in the power of sound with a guitar or trumpet, but all the instruments peacefully and harmoniously “made friends” in the overall sound image of the melody. At high frequencies, the UMZCH BB-2010, according to figurative audiophiles, sounds like “as if drawing a sound with a thin, thin brush.” These effects can be attributed to the difference in intermodulation distortion of the amplifiers.

The sound of the UMZCH Rotel RB 981 was similar to the sound of the NAD C352, with the exception of better work at low frequencies, yet UMZCH BB-2010 remained unrivaled in the clarity of AC control at low frequencies, as well as transparency, delicacy of sound at medium and high frequencies.

The most interesting in terms of understanding the mindset of audiophiles was the general opinion that, despite the superiority over these three UMZCH, they bring “warmth” to the sound, which makes it more pleasant, and UMZCH BB works smoothly, “it is neutral to the sound.”

The Japanese Dual CV1460 lost in sound immediately after being turned on in the most obvious way for everyone, and they did not waste time listening to it in detail. His Kr was in the range of 0.04 ... 0.07% at low power.

The main impressions from the comparison of amplifiers in general terms were completely identical: UMZCH BB was ahead of them in sound unconditionally and unambiguously. Therefore, further tests were considered unnecessary. As a result, friendship won, everyone got what they wanted: for a warm, intimate sound - Sugden, NAD and Rotel, and to hear what was recorded on the disc by the director - UMZCH BB-2010.

Personally, I like high-fidelity UMZCH with a light, clean, impeccable, noble sound, it effortlessly reproduces passages of any complexity. As my friend, an audiophile with great experience, put it, he works out the sounds of drum kits at low frequencies without options, like a press, at medium frequencies he sounds as if he does not exist, and at high frequencies he seems to paint the sound with a thin brush. For me, the non-irritating sound of UMZCH BB is associated with the ease of operation of cascades.

Literature

1. Sukhov I. UMZCH high fidelity. "Radio", 1989, No. 6, pp. 55-57; No. 7, pp. 57-61.

2. Ridiko L. UMZCH BB on a modern element base with a microcontroller control system. "Radiohobby", 2001, No. 5, pp. 52-57; No. 6, pp. 50-54; 2002, No. 2, pp. 53-56.

3. Ageev S. Superlinear UMZCH with deep environmental protection "Radio", 1999, No. 10 ... 12; "Radio", 2000, No. 1; 2; 4…6; 9…11.

4. Zuev. L. UMZCH with parallel environmental protection. "Radio", 2005, No. 2, p. 14.

5. Zhukovsky V. Why do we need the speed of UMZCH (or "UMZCH BB-2008"). "Radiohobby", 2008, No. 1, pp. 55-59; No. 2, pp. 49-55.

UMZCH VVS-2011 Ultimate version

UMZCH VVS-2011 Ultimate version author of the scheme Viktor Zhukovsky Krasnoarmeysk

Amplifier Specifications:
1. Big power: 150W / 8 ohm,
2. High linearity - 0.000.2 ... 0.000.3% at 20 kHz 100 W / 4 Ohm,
Full set of service nodes:
1. Maintain zero constant voltage,
2. AC wire resistance compensator,
3. Current protection,
4. DC voltage output protection,
5. Smooth start.

UMZCH VVS2011 scheme

The layout of printed circuit boards was carried out by a participant in many popular projects LepekhinV (Vladimir Lepekhin). It worked out very well).

UMZCH-VVS2011 board

ULF VVS-2011 amplifier board was designed for tunnel blowing (parallel to the radiator). The installation of transistors UN (voltage amplifier) ​​and VK (output stage) is somewhat difficult, because. installation / dismantling has to be done with a screwdriver through holes in the PCB with a diameter of about 6 mm. When access is open, the projection of transistors does not fall under the PP, it is much more convenient. I had to tweak the board a bit.

In the new software did not take into account one point- this is the convenience of setting the protection on the amplifier board:

C25 0.1n, R42 * 820 Ohm and R41 1k all smd elements are located on the soldering side, which is not very convenient when setting up, because it will be necessary to unscrew and fasten the bolts of the PCB on the racks and transistors to the radiators several times. Sentence: R42 * 820 consists of two smd resistors arranged in parallel, from here the proposal: we solder one smd resistor immediately, solder the other output resistor with a canopy to VT10, one pin to the base, the other to the emitter, we select it to the right one. Picked up, change the output to smd, for clarity:

UMZCH BB-2010 is a new development from the well-known line of amplifiers UMZCH BB (high fidelity). A number of technical solutions used were influenced by Ageev's work.

Specifications:

Harmonic distortion at 20,000 Hz: 0.001% (150 W/8 ohms)

-3dB Small Signal Bandwidth: 0 – 800000 Hz

Output voltage slew rate: 100V/µs

Signal-to-noise and signal-to-background ratio: 120 dB

Wiring diagram Air Force-2010

Due to the use of an op amp operating in a light mode, as well as the use of only stages with OK and OB in the voltage amplifier, covered by deep local OOS, UMZCH BB is highly linear even before the general OOS is covered. In the very first high-fidelity amplifier back in 1985, solutions were used that until then were used only in measuring technology: DC modes are supported by a separate service node, to reduce the level of interface distortions, the transient resistance of the AC switching relay contact group is covered by a common negative feedback, and a special node effectively compensates for the influence of the resistance of the AC cables on these distortions. The tradition has been preserved in UMZCH BB-2010, however, the general environmental protection also covers the resistance of the output low-pass filter.

In the vast majority of designs of other UMZCH, both professional and amateur, many of these solutions are still missing. At the same time, the high technical characteristics and audiophile advantages of the UMZCH BB are achieved by simple circuit solutions and a minimum of active elements. In fact, this is a relatively simple amplifier: one channel can be assembled slowly in a couple of days, and the setting consists only in setting the required quiescent current of the output transistors. Especially for beginner radio amateurs, a method has been developed for node-by-node, cascade-based performance testing and adjustment, using which you can guarantee to localize the places of possible errors and prevent their possible consequences even before the UMZCH is fully assembled. For all possible questions about this or similar amplifiers, there are detailed explanations, both on paper and on the Internet.

At the input of the amplifier, an R1C1 high-pass filter with a cutoff frequency of 1.6 Hz is provided, Fig. 1. But the efficiency of the mode stabilization device allows the amplifier to work with an input signal containing up to 400 mV of DC voltage. Therefore, C1 is excluded, which realizes the age-old audiophile dream of a path without capacitors and significantly improves the sound of the amplifier.

The capacitance of the capacitor C2 of the input low-pass filter R2C2 is chosen so that the cutoff frequency of the input low-pass filter, taking into account the output resistance of the preamplifier 500 Ohm -1 kOhm, is in the range from 120 to 200 kHz. The frequency correction circuit R3R5C3 is placed at the input of the op-amp DA1, which limits the band of processed harmonics and interference coming through the CUS circuit from the output side of the UMZCH to a band of 215 kHz at a level of -3 dB and increases the stability of the amplifier. This circuit makes it possible to reduce the difference signal above the cutoff frequency of the circuit and thus eliminate unnecessary overloading of the voltage amplifier by high-frequency interference, noise and harmonics, eliminating the possibility of dynamic intermodulation distortion (TIM; DIM).

Next, the signal is fed to the input of a low-noise operational amplifier with field-effect transistors at the input DA1. Many "claims" against the UMZCH BB are made by opponents regarding the use of an op-amp at the input, which allegedly degrades the sound quality and "steals the virtual depth" of the sound. In this regard, it is necessary to pay attention to some quite obvious features of the operation of the OS in the UMZCH VV.

Operational amplifiers of preamplifiers, post-DAC op-amps are forced to develop several volts of output voltage. Since the gain of the op-amp is small and ranges from 500 to 2000 times at 20 kHz, this indicates their operation with a relatively large difference signal voltage - from several hundred microvolts at low frequencies to several millivolts at 20 kHz and a high probability of introducing intermodulation distortions by the input stage of the op-amp. The output voltage of these op amps is equal to the output voltage of the last voltage amplification stage, usually made according to the OE scheme. An output voltage of several volts indicates the operation of this cascade with rather large input and output voltages, and as a result, it introduces distortions into the amplified signal. The op-amp is loaded with the resistance of the OOS circuit and the load connected in parallel, sometimes amounting to several kilo-ohms, which requires up to several milliamps from the output follower of the output current amplifier. Therefore, changes in the current of the output follower of the IC, the output stages of which consume a current of no more than 2 mA, are quite significant, which also indicates that they introduce distortions into the amplified signal. We see that the input stage, the voltage amplification stage and the output stage of the op-amp can introduce distortions.

But the high-fidelity amplifier circuitry, due to the high gain and input resistance of the transistor part of the voltage amplifier, provides very gentle operating conditions for the op-amp DA1. Judge for yourself. Even in the UMZCH, which has developed a rated output voltage of 50 V, the input differential stage of the op amp operates with differential voltage signals from 12 μV at frequencies of 500 Hz to 500 μV at a frequency of 20 kHz. The ratio of the high input overload capacity of the differential stage, made on field-effect transistors, and the meager voltage of the difference signal provides a high linearity of signal amplification. The output voltage of the op-amp does not exceed 300 mV. which indicates a low input voltage of the voltage amplification stage with a common emitter from the operational amplifier - up to 60 μV - and the linear mode of its operation. The output stage of the op-amp gives to the load about 100 kOhm from the side of the VT2 base an alternating current of not more than 3 μA. Consequently, the output stage of the op-amp also operates in an extremely lightweight mode, almost at idle. On a real musical signal, voltages and currents are, most of the time, an order of magnitude less than the given values.

From a comparison of the voltages of the difference and output signals, as well as the load current, it can be seen that, in general, the operational amplifier in the UMZCH BB operates hundreds of times easier, and, therefore, in a linear mode than the op-amp mode of preamplifiers and post-DAC op-amps of CD players that serve as sources signal for UMZCH with any depth of environmental protection, as well as without it at all. Consequently, the same op amp will introduce much less distortion as part of the UMZCH BB than in a single inclusion.

Occasionally there is an opinion that the distortions introduced by the cascade are ambiguously dependent on the voltage of the input signal. This is mistake. The dependence of the manifestation of the nonlinearity of the cascade on the voltage of the input signal may obey one law or another, but it is always unambiguous: an increase in this voltage never leads to a decrease in the introduced distortions, but only to an increase.

It is known that the level of distortion products attributable to a given frequency decreases in proportion to the depth of negative feedback for this frequency. The idle speed gain, up to the coverage of the feedback amplifier, at low frequencies cannot be measured due to the smallness of the input signal. According to calculations, the idle amplification developed up to the NOS coverage makes it possible to achieve an OOS depth of 104 dB at frequencies up to 500 Hz. Measurements for frequencies starting from 10 kHz show that the depth of the feedback at a frequency of 10 kHz reaches 80 dB, at a frequency of 20 kHz - 72 dB, at a frequency of 50 kHz - 62 dB and 40 dB - at a frequency of 200 kHz. Figure 2 shows the amplitude-frequency characteristics of UMZCH BB-2010 and, for comparison, similar in complexity.

High gain before coverage of the OOS is the main feature of the circuit design of VV amplifiers. Since the goal of all circuitry tricks is to achieve high linearity and high gain for maintaining deep feedback in the widest possible frequency band, this means that circuitry methods for improving amplifier parameters are exhausted by such structures. A further reduction in distortion can only be achieved by constructive measures aimed at reducing the pickup of harmonics of the output stage on the input circuits, especially on the inverting input circuit, the gain from which is maximum.

Another feature of the UMZCH BB circuitry is the current control of the output stage of the voltage amplifier. The input op-amp controls the voltage-to-current conversion stage, performed with OK and OB, and the received current is subtracted from the quiescent current of the stage, performed according to the OB circuit.

The use of a linearizing resistor R17 with a resistance of 1 kOhm in the differential stage VT1, VT2 on transistors of different structures with serial power increases the linearity of the conversion of the output voltage of the op-amp DA1 to the collector current VT2 by creating a local OOS with a depth of 40 dB. This can be seen from a comparison of the sum of the intrinsic resistances of the emitters VT1, VT2 - about 5 ohms each - with the resistance R17, or the sum of the thermal voltages VT1, VT2 - about 50 mV - with a voltage drop across the resistance R17, which is 5.2 - 5.6 V .

Amplifiers built according to the considered circuitry have a sharp, 40 dB per decade of frequency, gain decay above a frequency of 13 ... 16 kHz. The error signal, which is a distortion product, at frequencies above 20 kHz is two to three orders of magnitude smaller than the useful audio signal. This makes it possible to convert the linearity of the differential stage VT1, VT2, which is excessive at these frequencies, into an increase in the gain of the transistor part of the UN. Due to slight changes in the current of the differential stage VT1, VT2, when weak signals are amplified, its linearity does not deteriorate significantly with a decrease in the depth of the local OOS, but the operation of the op-amp DA1, on the operating mode of which the linearity of the entire amplifier depends on the operating mode of which at these frequencies, the gain margin will facilitate, since all voltages, The distortions that determine the distortions introduced by the operational amplifier, starting from the difference signal to the output signal, decrease in proportion to the gain in gain at a given frequency.

The phase advance correction circuits R18C13 and R19C16 were optimized in the simulator in order to reduce the difference voltage of the op-amp to frequencies of several megahertz. It was possible to increase the gain of UMZCH BB-2010 compared to UMZCH BB-2008 at frequencies of the order of several hundred kilohertz. Gain gain was 4 dB at 200 kHz, 6 dB at 300 kHz, 8.6 dB at 500 kHz, 10.5 dB at 800 kHz, 11 dB at 1 MHz, and 10 to 12 dB at frequencies above 2 MHz. This can be seen from the simulation results, Fig. 3, where the lower curve refers to the frequency response of the UMZCH BB-2008 lead correction circuit, and the upper one to UMZCH BB-2010.

VD7 protects the emitter junction VT1 from reverse voltage arising from the flow of recharging currents C13, C16 in the voltage limiting mode of the UMZCH output signal and the resulting limit voltages with a high rate of change at the output of the op-amp DA1.

The output stage of the voltage amplifier is made on a transistor VT3, connected according to a common base circuit, which excludes the penetration of a signal from the output circuits of the stage into the input circuits and increases its stability. The cascade with OB, loaded on the current generator on the VT5 transistor and the input impedance of the output stage, develops a high stable gain - up to 13,000 ... 15,000 times. The choice of the resistance of the resistor R24 ​​half the resistance of the resistor R26 guarantees the equality of the quiescent currents VT1, VT2 and VT3, VT5. R24, R26 provide local OOS that reduce the effect of the Earley effect - the change in p21e depending on the collector voltage and increase the initial linearity of the amplifier by 40 dB and 46 dB, respectively. The supply of the UN with a separate voltage, modulo 15 V higher than the voltage of the output stages, makes it possible to eliminate the effect of quasi-saturation of transistors VT3, VT5, which manifests itself in a decrease in n21e when the collector-base voltage drops below 7 V.

The three-stage output follower is assembled on bipolar transistors and does not require any special comments. Do not try to fight entropy by saving on the quiescent current of the output transistors. It should not be less than 250 mA; in the author's version - 320 mA.

Prior to the operation of the relay for switching on AC K1, the amplifier is covered by OOS1, implemented by turning on the divider R6R4. The accuracy of maintaining the resistance R6 and the consistency of these resistances in different channels is not essential, but to maintain the stability of the amplifier it is important that the resistance R6 is not much lower than the sum of the resistances R8 and R70. By actuating relay K1, the OOS1 is turned off and the OOS2 circuit, formed by R8R70C44 and R4, comes into operation, and covers the contact group K1.1, where R70C44 excludes the output low-pass filter R71L1 R72C47 from the OOS circuit at frequencies above 33 kHz. The frequency-dependent OOS R7C10 generates a decline in the frequency response of the UMZCH to the output low-pass filter at a frequency of 800 kHz at a level of -3 dB and provides a margin in depth of the OOS above this frequency. The frequency response decay at the AC terminals above the frequency of 280 kHz at a level of -3 dB is provided by the combined action of the R7C10 and the output low-pass filter R71L1 -R72C47.

The resonant properties of loudspeakers lead to the emission of damped sound vibrations by the diffuser, overtones after impulse action and the generation of its own voltage when the turns of the loudspeaker coil cross the magnetic field lines in the gap of the magnetic system. The damping coefficient shows how large the amplitude of the diffuser oscillations is and how quickly they decay when the AC is loaded as a generator on the impedance from the UMZCH. This coefficient is equal to the ratio of the AC resistance to the sum of the output resistance of the UMZCH, the transient resistance of the contact group of the AC switching relay, the resistance of the inductor coil of the output LPF usually wound with a wire of insufficient diameter, the transient resistance of the AC cable clamps and the resistance of the AC cables themselves.

In addition, the impedance of loudspeakers is non-linear. The flow of distorted currents through the wires of AC cables creates a voltage drop with a high degree of non-linear distortion, which is also subtracted from the undistorted output voltage of the amplifier. Therefore, the signal at the AC terminals is much more distorted than at the UMZCH output. These are the so-called interface distortions.

To reduce these distortions, compensation of all components of the total output impedance of the amplifier was applied. The own output resistance of the UMZCH, together with the contact resistance of the relay contacts and the resistance of the wire of the inductor of the output low-pass filter, is reduced by the action of a deep general OOS taken from the right output of L1. In addition, by connecting the right output of R70 to the “hot” AC terminal, you can easily compensate for the transient resistance of the AC cable clamp and the resistance of one of the AC wires, without fear of generating UMZCH due to phase shifts in the wires covered by the OOS.

The AC wire resistance compensation unit is made in the form of an inverting amplifier with Ky = -2 on the DA2, R10, C4, R11 and R9 op-amps. The input voltage for this amplifier is the voltage drop on the "cold" ("earth") wire of the speaker. Since its resistance is equal to the resistance of the "hot" wire of the AC cable, to compensate for the resistance of both wires, it is enough to double the voltage on the "cold" wire, invert it and through the resistor R9 with a resistance equal to the sum of the resistances R8 and R70 of the OOS circuit, apply to the inverting input of the op-amp DA1 . Then the output voltage of the UMZCH will increase by the sum of the voltage drops on the AC wires, which is equivalent to eliminating the influence of their resistance on the damping coefficient and the level of interface distortion at the AC terminals. Compensation for the drop in the resistance of the AC wires of the non-linear component of the back-EMF of loudspeakers is especially needed at the lower frequencies of the audio range. The signal voltage at the tweeter is limited by a resistor and capacitor connected in series with it. Their complex resistance is much greater than the resistance of the wires of the AC cable, so the compensation of this resistance at the RF is meaningless. Based on this, the integrating circuit R11C4 limits the operating frequency band of the compensator to 22 kHz.

Of particular note: the resistance of the "hot" wire of the AC cable can be compensated by covering it with a common OOS by connecting the right terminal of R70 with a special wire to the "hot" AC terminal. In this case, only the resistance of the "cold" AC wire will need to be compensated, and the gain of the wire resistance compensator must be reduced to the value Ku \u003d -1 by choosing the resistance of the resistor R10 equal to the resistance of the resistor R11.

The current protection unit prevents damage to the output transistors during short circuits in the load. Resistors R53 - R56 and R57 - R60 serve as a current sensor, which is quite enough. The amplifier output current flowing through these resistors creates a voltage drop that is applied to the divider R41R42. A voltage with a value greater than the threshold opens the transistor VT10, and its collector current opens the VT8 trigger cell VT8VT9. This cell goes into a steady state with open transistors and shunts the HL1VD8 circuit, reducing the current through the zener diode to zero and locking VT3. Discharging C21 with a small base current VT3 can take a few milliseconds. After the trigger cell is activated, the voltage on the lower plate of C23, charged by the voltage on the HL1 LED to 1.6 V, rises from the level of -7.2 V from the positive power rail of the UN to the level of -1.2 B1, the voltage on the upper plate of this capacitor also rises by 5 V. C21 is quickly discharged through the resistor R30 to C23, the transistor VT3 is closed. Meanwhile, VT6 opens and through R33, R36 opens VT7. VT7 shunts the zener diode VD9, discharges capacitor C22 through R31 and turns off transistor VT5. Not receiving a bias voltage, the output stage transistors are also locked.

Restoring the initial state of the trigger and turning on the UMZCH is done by pressing the button SA1 "Reset protection". C27 is charged by the VT9 collector current and shunts the VT8 base circuit, locking the trigger cell. If by this time the emergency has been eliminated and VT10 is locked, the cell goes into a state with stably closed transistors. VT6, VT7 are closed, a reference voltage is applied to the bases VT3, VT5 and the amplifier enters the operating mode. If the short circuit in the UMZCH load continues, the protection is activated again, even if the capacitor C27 is connected to SA1. The protection works so effectively that during the adjustment of the correction, the amplifier was de-energized several times for small soldering by touching the non-inverting input. The resulting self-excitation led to an increase in the current of the output transistors, and the protection turned off the amplifier. Although this crude method should not be offered as a rule, but due to current protection, it did not harm the output transistors.

Operation of AC cable resistance compensator

The efficiency of the UMZCH BB-2008 compensator was tested by the old audiophile method, by ear, by switching the compensator input between the compensating wire and the common wire of the amplifier. The improvement in sound was clearly noticeable, and the future owner was eager to get an amplifier, so no measurements of the effect of the compensator were carried out. The advantages of the cable-cutter scheme were so obvious that the compensator + integrator configuration was adopted as the standard assembly for installation in all developed amplifiers.

It's amazing how much unnecessary debate about the usefulness / uselessness of cable resistance compensation has flared up on the Internet. As usual, those who especially insisted on listening to a non-linear signal were complex and incomprehensible, the cost of it was exorbitant, and the installation was time-consuming ©. There were even suggestions that, since so much money is being spent on the amplifier itself, it’s a sin to save on the sacred, but you need to go the best, glamorous way that all civilized mankind goes and ... buy normal, human © super-expensive cables made of precious metals. To my great surprise, the statements of highly respected experts about the uselessness of the compensation unit at home, including those specialists who successfully use this unit in their amplifiers, added fuel to the fire. It is very unfortunate that many fellow radio amateurs were distrustful of reports about improving the sound quality at low and medium frequencies with the inclusion of a compensator, avoided this simple way to improve the operation of the UMZCH with all their might, than robbed themselves.

Little research has been done to document the truth. A number of frequencies were supplied from the GZ-118 generator to the UMZCH BB-2010 in the region of the AC resonant frequency, the voltage was controlled by an S1-117 oscilloscope, and Kr at the AC terminals was measured by INI C6-8, Fig. 4. Checking the effectiveness of wire resistance Resistor R1 is installed to avoid pickups at the input of the compensator when switching it between the control and common wires. The experiment used common and publicly available AC cables with a length of 3 m and a core cross section of 6 square meters. mm, as well as the GIGA FS Il speaker system with a frequency range of 25-22000 Hz, a nominal impedance of 8 ohms and a rated power of 90 W from Acoustic Kingdom.

Unfortunately, the circuitry of the harmonic signal amplifiers from the C6-8 composition provides for the use of high-capacity oxide capacitors in the environmental protection circuits. This causes the low-frequency noise of these capacitors to affect the resolution of the device at low frequencies, as a result of which its resolution at low frequencies deteriorates. When measuring Kr of a signal with a frequency of 25 Hz from GZ-118 directly from C6-8, the instrument readings dance around a value of 0.02%. It is not possible to get around this limitation using the GZ-118 generator notch filter in the case of measuring the compensator efficiency, because a number of discrete values ​​of the tuning frequencies of the 2T filter is limited at low frequencies by values ​​of 20, 60, 120, 200 Hz and does not allow measuring Kr at the frequencies of interest to us. Therefore, reluctantly, the level of 0.02% was taken as zero, the reference.

At a frequency of 20 Hz with a voltage at the AC terminals of 3 Vamps, which corresponds to an output power of 0.56 W into an 8 ohm load, Kr was 0.02% with the compensator on and 0.06% after it was turned off. At a voltage of 10 V amps, which corresponds to an output power of 6.25 W, the Kr value is 0.02% and 0.08%, respectively, at a voltage of 20 V amps and a power of 25 W - 0.016% and 0.11%, and at a voltage of 30 In the amplitude and power of 56 W - 0.02% and 0.13%.

Knowing the relaxed attitude of manufacturers of imported equipment to the values ​​​​of inscriptions regarding power, and also remembering the miraculous, after the adoption of Western standards, the transformation of an acoustic system with a subwoofer power of 30 W into , long-term power of more than 56 W was not supplied to AC.

At a frequency of 25 Hz at a power of 25 W, Kr was 0.02% and 0.12% with the compensation unit on / off, and at a power of 56 W - 0.02% and 0.15%.

At the same time, the necessity and effectiveness of covering the output LPF of the general OOS was checked. At a frequency of 25 Hz at a power of 56 W and connected in series to one of the wires of the AC cable of the output RL-RC low-pass filter, similar to that installed in the superlinear UMZCH, Kr with the compensator off reaches 0.18%. At a frequency of 30 Hz at a power of 56 W Kr 0.02% and 0.06% with the compensation unit on / off. At a frequency of 35 Hz at a power of 56 W, Kr is 0.02% and 0.04% with the compensation unit on / off. At frequencies of 40 and 90 Hz at a power of 56 W, Kr is 0.02% and 0.04% with the compensation unit on / off, and at a frequency of 60 Hz - 0.02% and 0.06%.

The conclusions are obvious. There is a presence of non-linear distortion of the signal at the AC terminals. The deterioration of the linearity of the signal at the AC terminals is clearly recorded with its inclusion through an uncompensated, uncovered OOS resistance of a low-pass filter containing 70 cm of a relatively thin wire. The dependence of the level of distortion on the power supplied to the AC suggests that it depends on the ratio of the signal power and the nominal power of the AC woofers. Distortions are most pronounced at frequencies near the resonant one. The back EMF generated by the speakers in response to the impact of an audio signal is shunted by the sum of the output resistance of the UMZCH and the resistance of the wires of the AC cable, so the level of distortion at the AC terminals directly depends on the resistance of these wires and the output impedance of the amplifier.

The cone of a poorly damped woofer itself emits overtones, and in addition, this loudspeaker generates a wide tail of harmonics and intermodulation distortion products that a midrange loudspeaker reproduces. This explains the deterioration of the sound at medium frequencies.

Despite the assumption of a zero Kr level of 0.02% due to the imperfection of the IRI, the effect of the cable resistance compensator on signal distortion on AC is clearly and unambiguously noted. It can be stated that the conclusions made after listening to the operation of the compensation unit on a musical signal and the results of instrumental measurements are in full agreement.

The improvement that is clearly audible when the cable cleaner is turned on can be explained by the fact that with the disappearance of distortion on the AC terminals, the midrange loudspeaker stops reproducing all this dirt. Apparently, therefore, by reducing or eliminating the reproduction of distortions by a mid-frequency loudspeaker, a two-cable AC connection circuit, the so-called. "biwiring", when the LF and MF-HF links are connected by different cables, has an advantage in sound compared to a single-cable circuit. However, since in a two-cable circuit the distorted signal at the terminals of the LF section of the AC does not disappear anywhere, this circuit loses to the option with a compensator in terms of the damping coefficient of the free vibrations of the cone of the low-frequency loudspeaker.

You can't deceive physics, and for a decent sound it is not enough to get brilliant performance at the output of the amplifier with an active load, but it is also necessary not to lose linearity after the signal is delivered to the speaker terminals. As part of a good amplifier, a compensator made according to one scheme or another is absolutely necessary.

Integrator

The effectiveness and possibility of reducing the error of the DA3 integrator was also tested. In UMZCH BB with op-amp TL071, the output DC voltage is in the range of 6 ... 9 mV, and it was not possible to reduce this voltage by including an additional resistor in the non-inverting input circuit.

The effect of low-frequency noise characteristic of a DC-input op-amp, due to the coverage of deep feedback through the frequency-dependent circuit R16R13C5C6, manifests itself in the form of an instability of the output voltage of a few millivolts, or -60 dB relative to the output voltage at rated output power, at frequencies below 1 Hz , not reproducible speakers.

On the Internet, it was mentioned the low resistance of the protective diodes VD1 ... VD4, which allegedly introduces an error into the operation of the integrator due to the formation of a divider (R16 + R13) / R VD2 | VD4 .. To check the reverse resistance of the protective diodes, a circuit was assembled Fig. 6. Here, the op amp DA1, connected according to the inverting amplifier circuit, is covered by the OOS through R2, its output voltage is proportional to the current in the circuit of the tested diode VD2 and the protective resistor R2 with a coefficient of 1 mV / nA, and the resistance of the R2VD2 circuit is with a coefficient of 1 mV / 15 GΩ . To eliminate the influence of the additive errors of the op-amp - bias voltage and input current on the results of measuring the diode leakage current, it is necessary to calculate only the difference between the intrinsic voltage at the output of the op-amp, measured without the diode under test, and the voltage at the output of the op-amp after its installation. In practice, a difference in the output voltages of the op-amp of several millivolts gives the reverse resistance of the diode of the order of ten to fifteen gigaohms at a reverse voltage of 15 V. It is obvious that the leakage current will not increase with a decrease in the voltage across the diode to a level of several millivolts, which is characteristic of the difference voltage of the op-amp of the integrator and compensator .

But the photoelectric effect inherent in diodes placed in a glass case really leads to a significant change in the output voltage of the UMZCH. When illuminated with an incandescent lamp of 60 W from a distance of 20 cm, the constant voltage at the output of the UMZCH increased to 20 ... 3O mV. Although it is unlikely that a similar level of illumination can be observed inside the amplifier case, a drop of paint applied to these diodes eliminated the dependence of the UMZCH modes on illumination. According to the simulation results, no drop in the frequency response of the UMZCH is observed even at a frequency of 1 millihertz. But the time constant R16R13C5C6 should not be reduced. The phases of the alternating voltages at the outputs of the integrator and the compensator are opposite, and with a decrease in the capacitance of the capacitors or the resistance of the resistors of the integrator, an increase in its output voltage can worsen the compensation of the resistance of the AC cables.

Amplifier sound comparison. The sound of the assembled amplifier was compared with the sound of several foreign industrial amplifiers. The source was a Cambridge Audio CD player, a pre-amplifier "" was used to build up and adjust the sound level of the terminal UMZCHs, the Sugden A21a and NAD C352 used standard controls.

The first to check was the legendary, outrageous and damn expensive English UMZCH "Sugden A21a", operating in class A with an output power of 25 watts. Remarkably, in the accompanying documentation for VCL, the British considered it good not to indicate the level of non-linear distortion. Say, it's not about distortions, but about spirituality. "Sugden A21a>" lost to UMZCH BB-2010 with comparable power both in terms of level and clarity, confidence, nobility of sound at low frequencies. This is not surprising, given the peculiarities of its circuitry: just a two-stage quasi-symmetric output follower on transistors of the same structure, assembled according to the circuitry of the 70s of the last century with a relatively high output resistance and an electrolytic capacitor switched on at the output that further increases the total output resistance - this is the last the solution itself degrades the sound of any amplifiers at low and medium frequencies. At medium and high frequencies, UMZCH BB showed higher detail, transparency and excellent stage elaboration, when singers and instruments could be clearly localized in sound. By the way, speaking of the correlation of objective measurement data and subjective impressions of the sound: in one of the magazine articles of Sugden's competitors, its Kr was determined at the level of 0.03% at a frequency of 10 kHz.

The next was also the English amplifier NAD С352. The general impression was the same: the pronounced "bucket" sound of the Englishman at low frequencies did not leave him any chances, while the work of the UMZCH BB was recognized as impeccable. Unlike NADa, whose sound was associated with thick bushes, wool, cotton wool, the sound of BB-2010 at medium and high frequencies made it possible to clearly distinguish the voices of performers in the general choir and instruments in the orchestra. In the work of NAD C352, the effect of better audibility of a more vociferous performer, a louder instrument, was clearly expressed. As the owner of the amplifier himself put it, in the sound of the UMZCH BB, the vocalists did not “shout out” to each other, and the violin did not fight in the power of sound with a guitar or trumpet, but all the instruments peacefully and harmoniously “made friends” in the overall sound image of the melody. At high frequencies, the UMZCH BB-2010, according to figurative audiophiles, sounds like “as if drawing a sound with a thin, thin brush.” These effects can be attributed to the difference in intermodulation distortion of the amplifiers.

The sound of the UMZCH Rotel RB 981 was similar to the sound of the NAD C352, with the exception of better performance at low frequencies, yet the UMZCH BB-2010 remained out of competition in the clarity of AC control at low frequencies, as well as transparency, delicacy of sound at medium and high frequencies.

The most interesting in terms of understanding the mindset of audiophiles was the general opinion that, despite the superiority over these three UMZCH, they bring “warmth” to the sound, which makes it more pleasant, and UMZCH BB works smoothly, “it is neutral to the sound.”

The Japanese Dual CV1460 lost in sound immediately after being turned on in the most obvious way for everyone, and they did not waste time listening to it in detail. His Kr was in the range of 0.04 ... 0.07% at low power.

The main impressions from the comparison of amplifiers in general terms were completely identical: UMZCH BB was ahead of them in sound unconditionally and unambiguously. Therefore, further tests were considered unnecessary. As a result, friendship won, everyone got what they wanted: for a warm, sincere sound - Sugden, NAD and Rotel, and to hear what was recorded on the disc by the director - UMZCH BB-2010.

Personally, I like high-fidelity UMZCH with a light, clean, impeccable, noble sound, it effortlessly reproduces passages of any complexity. As my friend, an audiophile with great experience, put it, he works out the sounds of drum kits at low frequencies without options, like a press, at medium frequencies he sounds as if he does not exist, and at high frequencies he seems to paint the sound with a thin brush. For me, the non-irritating sound of UMZCH BB is associated with the ease of operation of cascades.